Method and apparatus for the compression and/or transport and/or decompression of a digital signal

ABSTRACT

A method is described for compressing and/or transporting and/or decompressing a digital signal. The most significant bits of a sample of the digital signal are compressed and transported as a compressed transport sample. Methods are used to improve the compression and transport as hybrid DPCM and a dynamic shift of clip range of prediction errors. For every sample, not only a compressed transport sample but also a residual transport sample is transmitted. The residual transport sample is either equal to the least significant bits of the sample or equal to a substitution value which is a function of the clipping error from the compression of the most significant bits of the said sample. Apparatuses are described for compression and/or transport and/or decompression of a digitized television IF signal according to the method.

CROSS-REFERENCE TO RELATED APPLICATION

[0001] This application claims the benefit of European patentapplication No. 99870108.0 filed May 21, 1999, the contents thereofbeing incorporated hereinto by reference.

FIELD OF THE INVENTION

[0002] The present invention relates to methods and to apparatuses forthe compression and/or transport and/or decompression of digitalsignals, and for the compression and/or transport and/or decompressionof digitized television IF (Intermediate Frequency) signals inparticular. The primary field of application is that of cable televisionsystems.

DESCRIPTION OF RELATED ART

[0003] Methods and apparatuses are known for the transport of televisionsignals over analog communication paths as, e.g., cable televisionnetworks, whereby television signals are transported as analog IFsignals or modulated analog IF signals, the signals having a formataccording to one of the existing transmission standards as mentioned inITU-R Recommendation 470-2, “Television Systems”, 1986.

[0004] Methods are known to digitize IF signals. EP-0749237 andEP-0763899 of THOMSON multimedia S.A. describe such methods.

[0005] Methods are known to transport an uncompressed digitizedtelevision IF signal over a transport channel. In the Proceedings of theInternational TV Symposium, pp. 709-717, Montreux 1997, F. Van de Vyver,“BARCO Lynx: Digital Optical Solution for IF Transport of TelevisionSignals”, a method is described for transport of a digitized televisionIF signal over an optical fiber.

[0006] Methods are known to compress and transport digital signals,including digitized baseband audio and/or video signals. In televisionsystems with digital compression and transport, these methods are usedto compress audio and video separately before being multiplexed into onebit-stream of compressed data. For colored video, the bit-stream is inits turn a multiplex of occasional motion information and separatelycompressed components (e.g., luminance and color differences).

[0007] However, in case of a digitized television IF signal, which is asampled carrier modulated by the composite signal of audio and all videocomponents (and occasional additional data), all components arecompletely merged, making separate compression of each of themimpossible. Therefore, typical compression techniques for video andaudio cannot be used for the compression and/or transport of digitizedtelevision IF signals.

[0008] When a low implementation complexity and a lossless or nearlylossless compression are required, the compression ratio cannot be high.A possible choice is a split of the input in LSB's (Least SignificantBits*) which are not compressed and MSB's (Most Significant Bits) whichare compressed with predictive coding. In predictive coding, samples arecoded one by one. For each sample, first a predictor calculates aprediction of the coded sample based on previously coded samples. Thenthe prediction error, which is the difference between a sample and itsprediction, is transmitted to the receiver. There, the same prediction,calculated in the decoder, is added back to the received predictionerror. Consequently, the decoder is a recursive loop containing an adderand a predictor based on previously decoded samples. In order to reducethe bit-rate, the prediction errors can be quantized and/or clipped.This causes differences between the decoder output and the original.These differences are called coding errors. When there are coding errorsin the MSB's of a sample due to quantization or clipping of predictionerrors, transmission of the LSB's of the said sample is useless and theavailable channel bit-rate is not optimally used.

[0009] Predictive coding or Differential Pulse Code Modulation (DPCM) isoften combined with Variable Length Coding (VLC) or Huffman coding ofthe prediction errors. VLC coding significantly improves thecompression, but considerably increases the complexity. In order toguarantee compression in the case of DPCM without VLC, the transmittedprediction errors should be quantized and/or clipped to a range called“clip range”. However, in applications of lossless or nearly losslesscoding, quantization should be avoided and the prediction errors shouldbe as much as possible within the clip range.

[0010] In WO 83/03727, W. Kehler discloses a “Modulation and codingmethod with range prediction and reduced bit rate adapted to a signal”,which is an improved DPCM scheme that predicts a range called predictionrange instead of a single prediction value. Instead of the predictionerror, the position of the coded sample within the prediction range istransmitted. The prediction range can be used as clip range.

[0011] Predictive coding or DPCM is based on a recursive loop in thedecoder, and is therefore sensitive to transmission errors. Therobustness against transmission errors can be improved by Hybrid DPCM(HDPCM) as published by Van Buul in “Hybrid DPCM, a Combination of PCMand DPCM”, IEEE Trans. Commun., Vol. COM-26, No. 3, pp. 362-368, March1978. HDPCM has been applied by Van Buul to video only. After atransmission error, or when the decoder does not start decoding thetransmitted bit-stream from the beginning (i.e., after a random access),the predictions in the DPCM decoder are not the same as in the DPCMencoder. HDPCM forces the decoder predictions to converge to the encoderpredictions, as will be described in the enabling description of thefirst embodiment below. However, with HDPCM, when a decoder predictionconverges almost completely to the encoder prediction after atransmission error or after a random access, a big prediction error canbe decoded by the HDPCM decoder as a prediction error of nearly the sameabsolute value but with the opposite sign (described more in detailbelow), causing severe divergence between decoder and encoderpredictions. This is especially the case with digitized television IFsignals, where big prediction errors occur more often than in the caseof video signals.

SUMMARY OF THE INVENTION

[0012] It is an aim of the present invention to provide a method forprocessing a digital signal, so that it can be compressed andtransported over an existing transport channel featuring a known maximumbandwidth and/or a known maximum bit rate which is lower than thebandwidth respectively bit-rate of the said digital signal, therebyreducing the bit rate of the digital signal to be transmitted, andthereby keeping the distortions of the reproduced digital signal aftercompression, transport and decompression to a minimum. When the saiddigital signal is a digitized television IF signal, the visible andaudible distortions should remain minimum. It is an aim to keep theimplementation complexity of the method of this invention low and tokeep the disclosed compression lossless or nearly lossless.

[0013] This is achieved by a method for the transmission of a firstdigital signal from a first location over a transport channel to one ormore second locations where it is received as a second digital signal,which is substantially equal to the first digital signal. A sample ofthe first digital signal, represented by M bits, being the total of Nmost significant bits and M−N least significant bits, N being smallerthan or equal to M, is transported over the transport channel as atransport sample of a transport channel bit stream. The N mostsignificant bits are compressed to N−C bits, C being smaller than N andlarger than 0. The said N−C bits after compression are transported as a“compressed transport sample”. Next to the (N−C)-bit compressedtransport sample, there is a (M−N)-bit “residual transport sample” whichis transported over the transport channel.

[0014] The N−C bits of the compressed transport sample are obtainedthrough predictive coding of the N most significant bits of thecorresponding sample of the first digital signal, whereby to each sampleof the first digital signal corresponds at least a “prediction”, a“prediction error” and a “clipping error”. The prediction represents thepredicted N most significant bits of the sample of the first digitalsignal whereby the prediction is based on previously compressed samples.The prediction error represents the difference between the N mostsignificant bits of the sample of the first digital signal and the saidprediction. The clipping error represents the difference between theprediction error and a “clipped prediction error”, which is theprediction error clipped by a first clipper to a “clip range” [A . . .B], A and B being integers and B−A being equal to or smaller than2^((N−C)) ⁻¹. So, the clipped prediction error can be represented bymeans of N−C bits. The residual transport sample, represented by M−Nbits, is equal to the M−N least significant bits of the said sample ofthe first digital signal in the case that the said prediction errorcorresponding to the said sample of the first digital signal is in therange [(A+1) . . . (B-1)]. In the other case, it is equal to asubstitution value which is a function of on one hand the clipping errorcorresponding to the said sample of the first digital signal, and on theother hand the M−N least significant bits of the said sample of thefirst digital signal, whereby the output value of the said function ofthe clipping error and the least significant bits can be represented byM−N bits. It is preferred that, in the case that the prediction errorcorresponding to the said sample of the first digital signal is not inthe range [(A+1). (B−1)]. The said substitution value (CE) is obtainedby first taking the absolute value of the clipping error of the saidsample (S1) of the first digital signal (DS2), then clipping the saidabsolute value to a range which can be represented by P bits, with Pequal or less than M−N, and then using the remaining M−N−P bits for theoutput of a quantization to M−N−P levels of the M−N least significantbits of the said sample of the first digital signal.

[0015] The compressed transport sample can be the said clippedprediction error, but is by preference as in the range [A . . . B]wrapped around sum of the clipped prediction error and a mapped value ofthe prediction. The mapped value of the prediction is the predictionmapped on a range [D . . . E], E and D being integers and E−D beingequal to or smaller than 2^((N−C))−1. The prediction is by preferencemapped or quantized in a non-uniform way, such that the quantization isfine for prediction values corresponding to small input amplitudes andrough for prediction values corresponding to big amplitudes of the firstdigital signal.

[0016] Of the said clip range [A . . . B], B−A is by preference smallerthan 2^((N−C))−1 if the compressed transport sample is the in the range[A . . . B] wrapped around sum of clipped prediction error and mappedprediction.

[0017] The residual transport signal is by preference transported inPCM.

[0018] The clip range [A . . . B] can be fixed (being not shiftable) forall samples, but is by preference shiftable in every sample over a shiftsh, the shift sh in a sample being a function of one or more actualparameters of the said sample of the first digital signal, the cliprange then being [A+sh . . . B+sh]. When the to be transported digitalsignal is a digitized television IF signal, the said actual parametersare the estimated phase of the IF carrier and the estimated luminance ofthe video signal comprised in the television IF signal.

[0019] The aims of the invention are also achieved by other methodswhich combine only some of the method steps of the “preferred” method ofthe present invention described before, albeit with a lower gradecompression and/or transport and/or decompression of a digital signal.

[0020] A first method thus combining only some of the method steps ofthe preferred method of the present invention is a method for thetransmission of a first digital signal from a first location over atransport channel to one or more second locations where it is receivedas a second digital signal which is substantially equal to the firstdigital signal. A sample of the first digital signal represented by Nbits is transported over the transport channel as a transport samplerepresented by N−C bits, C being smaller than N and larger than 0. TheseN−C bits are obtained through predictive coding of the N bits of thesample of the first digital signal, whereby with each sample of thefirst digital signal corresponds at least a prediction representing thepredicted N bits of the sample of the first digital signal and aprediction error representing the difference between the N bits of thesample of the first digital signal and the said prediction. Theprediction is based on previously compressed samples. The transportsample is an in the range [A. B] wrapped around sum of the predictionerror clipped to a range named clip range [A. B], A and B being integersand B-A being equal to 2^((N−C)) ⁻¹, and a mapped value of theprediction which has, been mapped on a range [D . . . E], E and D beingintegers and E−D being equal to or smaller than ^((N−C))−1.

[0021] A second method thus combining only some of the method steps ofthe preferred method of the present invention is a method for thetransmission of a first digital signal from a first location over atransport channel to one or more second locations where it is receivedas a second digital signal which is substantially equal to the firstdigital signal. A sample of the first digital signal represented by Nbits is transported over the transport channel as a transport samplerepresented by N−C bits, C being smaller than N and larger than O. TheseN−C bits are obtained through predictive coding of the N bits of thesame of the first digital signal, whereby to each sample of the firstdigital signal corresponds at least a prediction representing thepredicted N bits of the sample of the first digital signal and aprediction error representing the difference between the N bits of thesample of the first digital signal and the said prediction. Theprediction is based on previously compressed samples. The transportsample is the prediction error clipped to a range named clip range,which can be represented by means of N−C bits, whereby the clip range isshiftable in function of one or more actual parameters of the firstdigital signal.

[0022] It is furthermore an aim of the present invention to provide anapparatus for the compression and/or transport and/or decompression of adigital signal, and of a digitized television IF signal in particular.

[0023] Transmitting apparatuses are provided wherein a digitizedtelevision IP signal is transformed into a transport channel bit-streamfor transmission of the said digitized television IF signal from a firstlocation to one or more second locations. Receiving apparatuses thatcorrespond to the provided transmitting apparatuses are also provided.These said receiving apparatuses transform a transport channelbit-stream, containing a first digitized television IF signal which hasbeen compressed according to the present invention, into a seconddigitized television IF signal by a method corresponding to the methodof compression of the said first digitized television IF signal.

[0024] A transmitting apparatus is provided, comprising a slitter, anencoder CPCM-core, an output for a the transport channel bit-stream, afirst location clipping detector, a first location substitutor, and anoccasional first location MSB corrector. Transmitting apparatuses areprovided wherein there is in addition a combination of a predictionmapper and a wrap-around adder and/or a combination of a phaselocked-loop, a luminance estimator, and a shift estimator.

[0025] A short description of the above listed parts of the providedtransmitting apparatuses is given below. For a more extended descriptionof these listed parts one is referred to the description of the firstembodiment of the present invention.

[0026] In the splitter, a sample of the digitized television IF signalis split into N most significant bits and M−N least significant bits. Inthe encoder DPCM-core, the N most significant bits of a sample of thesaid digitized television IF signal are compressed into an N−C-bit wordclipped prediction error. In the first location clipping detector, afirst location PCM-bit substitution control signal is generated,indicating what is to be transmitted as residual transport sample,either the M−N least significant bits of the sample of the firstdigitized television IF signal, or a substitution value being a functionof on one hand the clipping error corresponding to the said sample ofthe first digitized television IF signal, and on the other hand the M−Nleast significant bits of the said sample of the first digitizedtelevision IF signal. In the first location substitutor, the M−N leastsignificant bits are substituted by a substitution value according tothe first location PCM-bit substitution control signal. The occasionalfirst location MSB corrector adds to or subtracts from the sum ofprediction and the clipped prediction error, the output value of mappingby means of a second function of the received residual transport sample,according to the second location PCM-bit substitution control signal andthe sign signal.

[0027] In the prediction mapper, a mapped value is generated from anencoder prediction from the encoder DPCM-core or the sum of theprediction and the shift. In the wrap-around adder, a mapped predictionis added to a corresponding clipped prediction error, the sum then beingwrapped around to obtain a compressed transport sample.

[0028] In the phase-locked loop, the phase of the IF carrier of thedigitized television IF signal is estimated, based on a locally decodedtelevision IF signal from the encoder DPCM-core. In the luminanceestimator, the luminance of the video signal contained in the digitizedtelevision IF signal is estimated, based on a decoded television IFsignal and on the estimated phase of the IF carrier, resulting in anestimated luminance. In the shift estimator, a shift is estimated, basedon the estimated phase of the IF carrier and on the estimated luminance.

[0029] The combination of a phase-locked loop, a luminance estimator,and a shift estimator provides a shiftable clip range for the predictionerrors. In this case, the encoder DPCM-core contains a clipper thatclips the prediction error to a range that is shifted.

[0030] A receiving apparatus is provided, comprising an input for atransport channel bit stream, a decoder DPCM-core, a combiner, a secondlocation clipping detector, a second location substitutor, and a secondlocation MSB corrector. Receiving apparatuses are provided wherein thereis in addition a combination of a prediction mapper and a wrap-aroundsubtractor and/or a combination of a phase-locked loop, a luminanceestimator, and a shift estimator.

[0031] A short description of the above listed parts of the providedreceiving apparatuses is given below. A more extended description ofthese listed parts is given in the description of the first embodimentof the present invention.

[0032] In the decoder DPCM-core, the N−C bits of a sample of thecompressed transport stream are decompressed to N most significant bitsof an output sample. In the combiner, the M−N least significant bits andN most significant bits of a sample are combined to an output sample. Inthe second location clipping detector, a second location PCM-bitsubstitution control signal is generated, indicating what is to beselected as M−N least significant bits of the output sample, as well asa sign signal, being the sign bit of the clipping error. The secondlocation substitutor switches between the received M−N least significantbits from the residual bit stream and a fixed replacement according tothe second location PCM-bit substitution control signal. The secondlocation MSB corrector adds to or subtracts from the sum of predictionand received clipped prediction error, the output value of mapping bymeans of a second function of the received residual transport sample,according to the second location PCM-bit substitution control signal andthe sign signal.

[0033] Other transmitting and receiving apparatuses are provided whichare built up with only some of the parts used in the above describedprovided apparatuses and with equivalent mutual interconnections,however, offering a lower grade compression and/or transport and/ordecompression of a digital signal, a digitized television signal inparticular. One transmitting apparatus according to this inventionincludes at least an encoder DPCM core, an output for the transportchannel bit-stream, a prediction mapper, and a wrap-around adder, thecorresponding receiving apparatus also being provided and including atleast an input for the transport channel bit-stream, a decoderDPCM-core, a prediction mapper and a wrap-around subtractor.

[0034] Another transmitting apparatus provided by this inventionincludes at least an encoder CPCM core, an output for the transportchannel bit-stream, a phase-locked loop, a luminance estimator and ashift estimator, the corresponding receiving apparatus also beingprovided and including at least an input for the transport channelbit-stream, a decoder CPCM-core, a phase-locked loop, a luminanceestimator and shift estimator.

[0035] Still another transmitting apparatus according to this inventionincludes at least an encoder DPCM core, an output for the transportchannel bit-stream, a prediction mapper, a wrap-around adder, aphase-locked loop, a luminance estimator and a shift estimator, thecorresponding receiving apparatus also being provided and include atleast an input for the transport channel bit-stream, a decoderDPCM-core, a prediction mapper, a wrap-around subtractor, a phase-lockedloop, a luminance estimator and a shift estimator.

BRIEF DESCRIPTION OF THE DRAWINGS

[0036] The invention will now be described by means of the embodimentsrepresented in the draws.

[0037] In the drawings:

[0038]FIG. 1 shows a block diagram illustrating the transport of adigital signal from a first location over a transport channel to asecond location;

[0039]FIG. 2 shows a typical spectrum H(f) of a digitized television IFsignal;

[0040]FIG. 3 shows a sampled television IF signal with negativemodulation during an interval located around a vertical synchronizationpulse. For clarity, only the envelope of the maximum values and theenvelope of the minimum values are show, i.e., the maximum and minimumvalues in a sliding widow of 25 samples;

[0041]FIG. 4 shows a detail of FIG. 3 for the samples numbered from237280 to 237380. Illustrating sampled television IF signal withnegative modulation at a transition from a high to a low modulatingluminance;

[0042]FIG. 5 shows a block diagram illustrating the compression,transport and decompression of a digital signal according to a firstembodiment of the present invention;

[0043]FIG. 6 shows a block diagram illustrating an elementary but notcomplete implementation of the first, second and third embodiments ofthe present invention, whereby DPCM is implemented for transport of theN most significant bits;

[0044]FIG. 7 shows a block diagram illustrating a prior art method forcompression, transport and decompression of a digital signal using DPCM;

[0045]FIG. 8 shows a block diagram illustrating a DPCM encoder core anddecoder core of the present invention as implemented in the embodiments;

[0046]FIG. 9 shows a number of scales illustrating ranges of signalsthat are present in the embodiments of the present invention;

[0047]FIG. 10 shows a block diagram illustrating HDPCM (HybridDifferential Pulse Code Modulation) as implemented in the first andsecond embodiments;

[0048]FIG. 12 shows a block diagram illustrating a partialimplementation of the first embodiment of this invention, whereby DPCMis implemented for transport of the N most significant bits and PCM-bitsubstitution is carried out on the M−N least significant bits;

[0049]FIG. 12 shows a block diagram, illustrating a partialimplementation of the first embodiment of this invention, whereby HDPCMis implemented for transport of the N most significant bits and PCM-bitsubstitution is carried out on the M−N least significant bits;

[0050]FIG. 13 shows, for a television IF signal with negativemodulation, a transition from a high to a low luminance whereby thefirst sample after the transition coincides with a positive peak of theIF carrier;

[0051]FIG. 14 shows for a television IF signal with negative modulation,a transition from a low to a high luminance whereby the first sampleafter the transition coincides with a positive peak of the IF carrier;

[0052]FIG. 15 shows a block diagram, illustrating the addition of adynamic clip range shifting to the elementary implementation of thefirst, second and third embodiments of the present invention, the blockdiagram corresponding to the third embodiment of the present invention;

[0053]FIG. 16 shows a block diagram of an IF carrier estimation, being aphase-locked loop (PLL);

[0054]FIG. 17 shows a block diagram of a luminance estimator (LUE);

[0055]FIG. 18 shows a block diagram of a shift estimator (SHE);

[0056]FIG. 19 shows a block diagram illustrating an implementation of adynamic clip range shifting;

[0057]FIG. 20 shows several time diagrams to illustrate the time delaybetween a sample and a corresponding calculated shift amount;

[0058]FIG. 21 shows a block diagram, corresponding to the secondembodiment of the present invention; and

[0059]FIG. 22 shows a block diagram of a prediction error clipper(PEC1).

[0060]FIG. 23 shows a block diagram illustrating an alternative partialimplementation of the first embodiment of this invention, whereby DPCMis implemented for transport of the N most significant bits and PCM-bitsubstitution is carried out on the M−N least significant bits;

[0061] In the drawings, same elements are represented by the samereferences.

ENABLING DESCRIPTION OF PREFERRED EMBODIMENTS

[0062] First Preferred Embodiment

[0063]FIG. 1 shows a first embodiment of the present invention which isa method for the transport of a first digitized television IF signal DS1over a transport channel TC from a first location FL to a secondlocation SL, where it is received as a second digitized television IFsignal DS2. The bit-rate of the digitized television IF signal DS1 isfitted in the available channel bit rate, according to the presentinvention. The first digitized television IF signal DS1 is a modulatedtelevision signal at an IF according to one of the existing transmissionstandards as mentioned in ITU-R Recommendation 470-2, “TelevisionSystems”, 1986, and modulated at an IF carrier frequency.

[0064] At the first location FL, the first digitized television IFsignal DS1 is transformed in a transmitter TRA into a transport channelbit-stream TCBS. The transport channel bit-stream TCBS is transmittedover the transport channel TC from the first location FL to the secondlocation SL. At the second location SL, the received transport channelbit-stream TCBS is transformed in a receiver REC into a second digitizedtelevision IF signal DS2.

[0065] The first digitized television IF signal DS21 may optionally bethe output of an analog-to-digital converter AD with a first analogtelevision IF signal AS1 at its input. The second digitized televisionIF signal DS2 may optionally be the input of a digital-to-analogconverter DA with a second analog television IF signal AS2 at itsoutput.

[0066] As it is an aim of the invention to keep the visible and audibledistortions to a minimum when reproducing the transported digitizedtelevision IF signal, the second digitized IF signal DS2 should besubstantially equal to the first digitized television IF signal DS1.

[0067] For the first embodiment, a television IF signal according to theNTSC M standard is considered. The IF frequency is 45.75 MHz.

[0068] The first digitized television IF signal DS1 is sampled at asampling rate of 16.2 MHz with 11 bits per sample. The clock thatindicates the sampling time points is called the “sampling clock”.

[0069] The bit rate of the first digitized television IF signal DS1 is16.2*11 or 178.2 Mbits/s. Over the transport channel TC (e.g. SONET OC-3or SDH STM-1), signal transport with a payload bit-rate up to 149.76Mbits/s is available.

[0070]FIG. 2 shows a typical spectrum of a digitized television IFsignal, whereby a NTSC M baseband television signal is modulated to anIF carrier with a frequency of 45.75 MHz, sampled at 16.2 MHz.

[0071] The horizontal axis f is a frequency scale between 0 MHz and 8.1MHz, and the vertical axis H(f) is an amplitude scale between −50 dB and+60 dB.

[0072]FIG. 3 shows a sampled television IF signal during an intervallocated around a vertical synchronization pulse, the sampling frequencybeing 16.2 MHz. The horizontal axis SN (sample number) is a time scalewith numbers of consecutive samples from 210000 to 245000 as units, andthe vertical axis SV (sample value) is an amplitude scale representingthe amplitude of the samples of the sampled television IF signal aftersampling with 11 bits per sample. Due to the negative modulation, anamplitude peak which corresponds to a horizontal sync pulse is visibleapproximately every 1000 samples. FIG. 4 shows a detail of FIG. 3 forthe samples numbered from 237280 to 237380, illustrating the sampledtelevision IF signal at a transition from a high to a low modulatingluminance.

[0073] The above mentioned parameters of the first digitized televisionIF signal DS1 to be transported (NTSC M as transmission standard, 45.75MHz as IF carrier frequency, 16.2 MHz as sampling rate, 11 bits persample), and the above mentioned specifications of the transport channelare choices made to describe this first embodiment of the presentinvention. Other typical parameters and specifications known to a personskilled in the art can easily replace them. The intention is to coverall modifications, equivalents and alternatives falling within thespirit of the present invention.

[0074]FIG. 5 shows a block diagram of a full implementation of the firstembodiment of the present invention. The description of this embodimentis broken down into smaller sections and is built up step-by-step,whereby the sections of the description correspond to parts wherein thisembodiment and the method of the invention can be subdivided, and isreferred to additional figures. As will be seen from the descriptionbelow, the consecutively described parts of this embodiment and theirinterconnections from an elementary implementation to a fullimplementation correspond to a gradually improving method for obtainingthe aims of the invention. The parts of this embodiment are thereforenamed “elementary implementation” and “improving parts”. Some of theseimproving parts can operate independent from other improving parts.Consequently, a previously described improving part is not necessarilyneeded for the operation of an improving part described further. Some ofthe improving parts contain one or more improvements themselves (orextra-improving parts) which are also introduced step-by step, and/orone or more divisional parts which draw up the improving part. For thesake of clarity, an overview is given here of how the description ofthis first embodiment of the invention is built-up. After a descriptionof the “elementary implementation”, “HDPCM” is described as a firstimproving part. The description of HDPCM includes a description of HDPCMitself followed by descriptions of two HDPCM extra-improving parts forHDPCM, namely “reduction of the clip range” and “non-uniform mapping”.Then follows the description of a second improving part named “PCM-bitsubstitution” which is not subdivided. Then follows a description of athird improving part named “Dynamic clip range shifting” comprisingthree divisions which are, after an introduction, describedconsecutively and which are named “IF carrier estimation”, “luminanceestimation” and “shift calculation”. The description of the shiftcalculation comprises the description of a further improvement called“dejitter”, and an extra-improving part to the combination of dynamicclip range shifting and HDPCM.

[0075]FIG. 5 shows the path of a sample S1 of the first digitizedtelevision IF signal from the first location FL, where it has beentransformed in a sample of the transport channel bit stream, over thetransport channel TC to the second location SL, where it is transformedinto a sample S2 of the second digitized television IF signal. When thedescription of this embodiment mentions from here on a sample, it meansthe sample S1 or a transformed version of that same sample S1.

[0076] The parts on the left side of FIG. 5 belong to the first locationFL and correspond to the transmitter TRA in FIG. 1. The parts on theright side of FIG. 5 belong to the second location SL and correspond tothe receiver REC in FIG. 1.

[0077]FIG. 6 shows the elementary implementation of this embodiment,omitting all improving parts described later.

[0078] A sample S1 of the first digitized television IF signal, having abit-resolution of M bits (11 in this embodiment), is first split at thefirst location FL with a splitter SP into N-MSB's (Most SignificantBits) and M−N LSB's (Least Significant Bits). The value N is bypreference 8, and M−N is consequently 3. On FIG. 5, FIG. 6 and otherfigures, the bit resolution of signals passing through some of theinterconnections is marked by means of a slash and the bit-resolution(e.g. “/M”, “/N” or “/N−C”).

[0079] The M−N LSB's, having usually a noisy character, are difficult tocompress and are therefore, as far as they are not replaced by otherbits (see below), by preference transmitted uncompressed in PCM (PulseCode Modulated). Therefore they are further also called “PCM-bits”.

[0080] The N MSB's are compressed to N−C bits in a further describedencoder DPCM-core D1 using DPCM (Differential Pulse Code Modulation) andtransmitted as an N−C-bit word. Therefore, the said N MSB's aresometimes called “DPCM-bits”, whereas the bits of the N−C-bit word aresometimes called the “compressed DPCM-bits”. In this embodiment, C ischosen to be 2, N−C consequently being 6. However, any other choice maybe made for C.

[0081] The transport channel bit-stream TCBS of FIG. 1 thus consists ofa compressed bit-stream (of compressed DPCM-bits) and a residualbit-stream (of PCM-bits). In FIGS. 5 and 6, the compressed bit-stream isrepresented by a compressed transport sample CTS, and the residualbit-stream is represented by a residual transport sample RTS, both beingtransformed versions of a part of the sample S1.

[0082] At the second location SL, the N−C bits of a received N−C-bitword are decompressed to N MSB's in a decoder DPCM-core D2 describedbelow. The N decompressed MSB's are combined in a combiner CB with thereceived corresponding M−N LSB's of the same sample, as far as theseLSB's have not been replaced by other bits (see below), to form anM-bits sample S2 of the second television IF signal.

[0083] The split of the M-bit samples S1 of the first digitizedtelevision IF signal into two parts has the advantage that the width ofthe DPCM-loop can be smaller than M bits (smaller than 11 bits for thisembodiment), which reduces the complexity of the hardware for thecompression and which permits a sufficiently fast operation of theDPCM-loop.

[0084] The DPCM part of this embodiment, consisting of two DPCM-cores,being an encoder DPCM-core D1 in an encoder at the first location FL anda decoder DPCM-core D2 in a decoder at the second location SL, is nowfurther explained. By the encoder is meant the part of the transmitterwhere the DPCM-bits of a sample S1 are transformed or encoded to acompressed transport sample CTS, the encoder DPCM-core D1 being a partor the whole of it. By the decoder is meant the part of the receiverwhere a received compressed transport sample CTS is transformed ordecoded to DPCM-bits of a sample S2, the decoder DPCM-core D2 thus beinga part or the whole of it.

[0085]FIG. 7 shows a prior art DPCM configuration for transmitting andreceiving a digital signal. A sample X of the digital signal to betransported over a transport channel TC is encoded before beingtransmitted as a compressed transport sample CTS making use of apredictor PR1, a quantizer and clipper Q&CL, an adder ADD11, asubtractor SUB11 and a variable length coder VLC, together forming anencoder. The received transport sample CTS is decoded to become areconstructed sample {circumflex over (x)}_(dec) of the digital signalto be transported, making use of a variable length decoder VLD, apredictor PR2 and an adder ADD21, together forming a decoder.

[0086] In both encoder and decoder, the same prediction is made frompreviously transmitted and decoded samples of the digital signal to betransported.

[0087] The encoder contains two recursive loops LP11 and LP12, whereasthe decoder contains only one recursive loop LP21. The loop LP21corresponds to loop LP12, and therefore, LP12 is called the “localdecoder in the encoder”.

[0088] In the encoder of the prior art DPCM configuration, a prediction{circumflex over (x)}_(enc) of a sample x, made by the predictor PR1from the previously locally decoded samples {tilde over (x)}_(enc), issubtracted in SUB11 from the sample x, resulting in a prediction errore_(enc). The prediction error e_(enc) is quantized and/or clipped in thequantizer and clipper Q&CL to become a clipped prediction error(e_(enc))_(c). This quantizing and/or clipping is carried out in orderto limit the number of possible code words to be transmitted. Theclipped prediction error (e_(enc))_(c) is added in ADD11 to theprediction {circumflex over (x)}_(enc), resulting in a reconstructedinput sample {tilde over (x)}_(enc) (being a locally decoded sample inthe encoder), which is the input of the predictor PR1 for making aprediction of future samples.

[0089] The quantized and clipped prediction error (e_(enc))_(c) is inputto the optional variable length coder VLC of which the output is thecompressed transport sample CTS. As prediction errors usually have ahistogram that is centered around zero, variable-length coding cansignificantly reduce the bit-rate of the prediction errors.

[0090] In the decoder of the prior art DPCM configuration, the receivedcompressed transport sample CTS is applied to an optional variablelength decoder VLD of which the output is a clipped prediction error(e_(dec))_(c) of the decoder. The clipped prediction error (e_(dec))_(c)is applied to a recursive decoder loop LP21 formed by the adder ADD21and the predictor PR2. The prediction error (e_(dec))_(c) is added inADD21 to a prediction {circumflex over (x)}_(dec) made by the predictorPR2 from the previously decoded sample, resulting in the reconstructedsample {tilde over (x)}_(dec).

[0091] If there are no transmission errors and when the encoder anddecoder start synchronously, the outputs of both loops LP12 and LP21,respectively {tilde over (x)}_(enc) and {tilde over (x)}_(dec), areequal.

[0092]FIG. 8 shows the DPCM part of the first embodiment of the presentinvention, being an adapted version of the prior art DPCM configurationshown in FIG. 7. In order to limit the complexity of the implementation,by preference no variable length coding is implemented. Withvariable-length coding, a difficult word synchronization has to beimplemented using synchronization words, a bit-rate and buffer control,stuffing and an exception handling for cases of buffer overflow. If novariable length coding is implemented, the clipped and quantizedprediction errors are coded with words of fixed length.

[0093] A prediction error e_(enc) is only clipped; therefore thequantizer and clipper Q&CL is replaced by a prediction error clipperPEC1. Quantization is in this preferred embodiment not implementedbecause it is an aim of the invention to have a lossless or nearlossless compression.

[0094] Due to the fixed length coding, the number of available codewords in the codebook is limited. Therefore, in the encoder theprediction errors e_(enc) are clipped to a range [A . . . B], furtheralso called the clip range. A and B are integers with B−A+1≦2^((N−C)),which allows a (N−C)-bit code word for every possible value of theclipped prediction error.

[0095] For this embodiment, N−C is 6 bits. The preferred range ofprediction errors for this embodiment is [−32 . . . 31]. However, forreasons explained below, the range of prediction errors will be reducedby preference to [−31 . . . +30]. In some cases, the clip range will beshifted over an amount “sh” to the range [A+sh . . . B+sh], with shvarying from sample to sample as will be described below. A preferredkernel for compression of digitized television IF-signals of thisembodiment is a DPCM-loop using an 8-tap linear prediction (LPC). Thecoefficients of the linear prediction are optimized for the statisticalcharacteristics of a representative number of television IF signals.FIG. 9 shows a number of scales illustrating the ranges of signals whichare present in this embodiment of the present invention for which Nequals 8 and C equals 2. Scale SCL1 is for the N MSB's X, scales SCL2and SCL6 for the predictions {circumflex over (x)}, scale SCL3 for thenot clipped prediction errors e, and scale SCL4 for the clippedprediction errors e_(c); on top of each scale, the bit-resolution ismentioned. The scales SCL6, SCL7, SCL8 and SCL9 are described below.

[0096] First Improving Part, Named “HDPCM”

[0097] In order to improve the transport of samples, or morespecifically, to improve the robustness against transmission errors, andto allow a random access of the decoder to the transported bit-stream ofcompressed data so that the decoder can start decoding at any time afterstart-up of the encoder without waiting for start codes, the abovedescribed encoder and decoder are by preference extended to HybridDifferential Pulse Code Modulation or HDPCM. This HDPCM is an improvingpart. HDPCM is described in IEEE Transactions on CommunicationTechnology, Vol. COM-26, No. 3, pp. 362-368, March 1978, Van Buul,“Hybrid DPCM, a combination of PCM and DPCM”. FIG. 10 shows theextension of the above described encoder and decoder DPCM-cores D1 andD2 with prediction mappers PM1 and PM2 respectively, in order to obtainHDPCM. With HDPCM, both the encoder and the decoder map theirpredictions on a value using the same prediction mapping. In a mapping,each possible input value has one corresponding output value. Thismapping can be considered as a quantization of the prediction. Thismapping preserves the ordering of the prediction values. This means thatwhen the encoder prediction {circumflex over (x)}_(enc) is higher thanthe decoder prediction ê_(dec), because of a previous transmission erroror because of a random access of the decoder, the mapped value of theencoder prediction m(ê_(enc)) is also higher than or equal to the mappedvalue of the decoder prediction m({circumflex over (x)}_(dec))

[0098] if {circumflex over (x)}_(enc)>{circumflex over (x)}_(dec),

[0099] then m({circumflex over (x)}_(enc))≧m({circumflex over(x)}_(dec))

[0100] The range of the mapping output is [D . . . E] with D and E beingintegers and E−D being equal to or less than 2^((N−C)). For thisembodiment with N−C=6, [D . . . E] is by preference [−32 . . . +31].

[0101] For a given sample in a HDPCM coding system, it is not theprediction error that is transmitted, but the sum of the clippedprediction error (e_(enc))_(c) and the mapped encoder predictionm({circumflex over (x)}_(enc)) In the block diagram of HDPCM in FIG. 10,the adder ADD12 in the encoder makes the sum of (e_(enc))_(c) andm({circumflex over (x)}_(enc)). At the receiver, the mapped predictionof the decoder m({circumflex over (x)}_(dec)) is then subtracted fromthe received sum of prediction error and mapped encoder prediction,resulting in a decoder output (e_(dec))_(c) of the received sample(subtractor SUB21 in the decoder in FIG. 10).

[0102] For the above addition and subtraction of prediction error andmapped prediction, the results are wrapped-around. In the presentembodiment with N−C or 6 DPCM bits, this wrap-around is a kind ofmodulo-operation which is mathematically described by the followingequations, where “mod” means the modulo operation and where y is aninteger number:

wr(y)=((y+2⁽⁶⁻¹⁾) mod 2⁶)−2⁽⁶⁻¹⁾ or

wr(y)=((y+32)mod 64)−32

[0103] Consequently, wr(+32)=−32, wr(+33)=−31, and so on, andwr(−33)=+31, wr(−34)=+30, and so on.

[0104] In FIG. 9, scale SCL5 illustrates the range of the mappedpredictions m({circumflex over (x)}), scale SCL7 the range of the sumsof a clipped prediction error e_(c) and a mapped predictionm({circumflex over (x)}), and scale SCL8 the range of the wrapped aroundsums of a clipped prediction error e_(c) and a mapped predictionm({circumflex over (x)}); on top of each scale, the bit-resolution ismentioned. The wrap-around is illustrated by means of dashed linesbetween the scales SCL7 and SCL8.

[0105] Because of the wrap-around, the number of possible code wordsremains the same for HDPCM as for DPCM. For this described embodimentwith N−C or 6 DPCM-bits, the number of code words for DPCM or HDPCM is2^((N−C)) or 26 or 64.

[0106] The clipped prediction error (e_(dec))_(c) which is the input tothe DPCM decoder (see FIG. 10) is given by:

(e _(dec))_(c) =wr(wr((e _(enc))_(c) +m({circumflex over(x)}_(enc)))−m({circumflex over (x)}_(dec)))

[0107] Because of the nature of the wrap-around, this can be simplifiedto:

(e _(dec))_(c=) wr((e _(enc))_(c) +m({circumflex over (x)}_(enc))−m({circumflex over (x)} _(dec))

[0108] When encoder and decoder have the same prediction for a givensample, i.e., {circumflex over (x)}_(dec)={circumflex over (x)}_(enc),the mapped decoder prediction m({circumflex over (x)}_(dec)) is equal tothe mapped encoder prediction m({circumflex over (x)}_(enc)), and thus(e_(dec))_(c)=(e_(enc))_(c).

[0109] When encoder and decoder don't have the same prediction, and if

−32≦(e _(enc))_(c) +m({circumflex over (x)}_(enc))−m({circumflex over(x)} _(dec))≦+31

[0110] then:

(e _(dec))_(c)=(e _(enc))_(c) +m({circumflex over (x)}_(enc))−m({circumflex over (x)} _(dec))

[0111] In this case, which happens almost all the time, it is as if thewrap-around in the decoder neutralizes the wrap-around in the encoder.

[0112] So, if the prediction {circumflex over (x)}_(dec) in the decoderis smaller than the prediction {circumflex over (x)}_(enc) in theencoder because of a transmission error in a previous sample or a randomaccess, the clipped prediction error (e_(dec))_(c) in the decoder willbe higher than or equal to the clipped prediction error (e_(enc))_(c) inthe encoder. If the clipped prediction error (e_(dec))_(c) in thedecoder is higher than the clipped prediction error (e_(enc))_(c) in theencoder, the too low prediction ê_(dec) of the decoder will becorrected. Indeed, {circumflex over (x)}_(dec)+(e_(dec))_(c), which isthe decoder output, will deviate less from {circumflex over(x)}_(enc)+(e_(enc))_(c) than {circumflex over (x)}_(dec) deviates from{circumflex over (x)}_(enc).

[0113] First Extra-Improving Part to HDPCM, Named “Reduction of the ClipRange”

[0114] When encoder and decoder don't have the same prediction, and if

(e _(enc))_(c) +m({circumflex over (x)} _(enc))−m({circumflex over (x)}_(dec))<−32 or

+31<(e _(enc))_(c) +m({circumflex over (x)}_(enc))−m({circumflex over(x)} _(dec))

then (e _(dec))_(c)=(e _(enc))_(c) +m({circumflex over (x)}_(enc))−m({circumflex over (x)} _(dec))+n*64

[0115] with n being an integer not equal to zero. In this case, thecombination of wrap-around in the encoder adder ADD12 and decodersubtractor SUB21 will cause a severe deviation between (e_(dec))_(c) and(e_(enc))_(c). This case fortunately seldom happens. If it happens, itis usually when the difference between the predictions {circumflex over(x)}_(enc) and {circumflex over (x)}_(dec) of resp. encoder and decoderare small and when at the same time the absolute prediction error isnear to its maximum value. A small difference between the predictions ofencoder and decoder occurs usually just before the completeresynchronization of the decoder with respect to the encoder. A largeprediction error occurs at sudden changes or non-stationarities of thevideo signal. In this embodiment with clipping to 6 bits, the extremeprediction errors are +31 and −32. If the clipped prediction error(e_(enc))_(c) is for instance +31 and at th same instant, the mappedencoder prediction (m({circumflex over (x)}_(enc))) is +1 higher thanthe mapped decoder prediction (m({circumflex over (x)}_(dec))), thedecoder will end up with a prediction error equal to +31+1 or +32 andthis will be wrapped around to −32. So, the prediction error(e_(dec))_(c) in the decoder has almost the same absolute value as theprediction error (e_(enc))_(c) in the encoder, but the sign is theopposite. This looks as if there is a transmission error of 31−(−32) or+63. To avoid such large errors in the output sufficiently, theprediction error clip range is by preference reduced, in this embodimentfrom [−22 . . . +31] to [−31 . . . +30], while the adder in the HDPCMencoder and the subtractor in the HDPCM decoder keep their wrap-aroundfor the range [−32 . . . +31].

[0116] The reduction of the clip range as described above causes itinstationarities in the video information only errors that are almostimperceptible. However it has the advantage that the decoder correctsmuch better than deviation of the decoder from the encoder aftertransmission errors or after a random access.

[0117] Second Extra-Improving Part TO HEPCM, Named “Non-Uniform Mapping”

[0118] HDPCM more quickly corrects the deviation of the decoder from theencoder when the transported signal has a higher amplitude. Withnegative modulation, the correction happens more quickly for sync pulsesthan for bright image parts. If this transported image is for instance acompletely white field, a transmission error typically lasts until theend of the horizontal line.

[0119] The error correcting efficiency of the applied HDPCM can beequalized between dark and bright image areas by an appropriatenon-uniform mapping of the prediction. For this embodiment withfixed-length-coding and 6 DCPM-bits per sample, there are 64 differentcode words. So, for HDPCM the predictions have to be mapped on 64values. It is known by practice that the prediction range isapproximately [−128. +127], some so-called “outliers” lying outside thisrange. Such a range [−128 . . . +127] could be mapped on 64 values by adivision by 4. With negative modulation, a typical range of predictionscorresponding to bright image parts is [−16 . . . +15]. If the HDPCMmapping would be a division by 4, only about 8 different output levelsof the prediction mapping would be used in bright areas. A typical rangeof predictions corresponding to dark image parts is [−64 . . . +63].With a prediction mapping which is a division by 4, about 32 differentoutput levels of the prediction mapping would be used in dark areas.Considering that the probability that transmission errors are correctedrises for an increasing probability that mapped predictions aredifferent in encoder and decoder, a division by 4 leads to a discrepancyof error correction between bright and dark image areas. Therefore, itis preferred to choose for this embodiment the following non-uniformmapping: one-to-one in the range [−16. +15], a division by 5 in theranges [−96 . . . −17] and [+16 . . . +95], and mapping to −32 forvalues less than −96 and to +31 for values higher than +95. Thispreferred mapping is illustrated in FIG. 9 by means of the dashed linesbetween the scales SCL6 and SCL5. With this non-uniform mapping, about32 mapping levels are used in bright areas, whereas 52 levels are usedin dark areas. This is a ratio of 52/32 for non-uniform mapping insteadof 32/8 for a uniform mapping based on a division by 4.

[0120] The above described strong error correction for bright imageparts through a non-uniform mapping is advantageous in order to obtain asufficiently correct estimation of the IF carrier phase which isdescribed below.

[0121] Second Improving Part, Named “PCM-Bit Substitution”

[0122] After the above description of the transport of the MSB's orDPCM-bits applying DPCM (or HDPCM), now follows a description of thetransport of the LSB's or PCM-bits.

[0123] When the MSB's are coded in DPCM (or HDPCM) and the LSB's arecoded in PCM, the LSB's loose their relevance when there is a codingerror in the DPCM (or HDPCM) output. A coding error in this embodimentwithout quantization is caused by clipping the prediction error. Whenthe prediction error has been clipped, the clipped prediction error isequal to a clipping value, so for this embodiment with 6 DPCM-bits, thisis equal to −31 or +30 (or −32 or +31 when the clip range has not beenreduced). When the decoder of this embodiment receives a predictionerror of −31 or +30, the decoder knows without the need for overheadinformation that clipping has most probably happened. It is only whenthe unclipped prediction error was exactly −31 or +30 that no clippinghas happened.

[0124] Consequently, when the transmitted clipped prediction error is+30 or −31, the PCM-bits may be substituted by the amount of theabsolute clipping error. This substitution is called a PCM-bitsubstitution and is an improving part of the invention. With 3 (=M−N)PCM-bits available, an absolute error of up to 7 can be transported asan unsigned integer. It is not necessary to transport a sign, becausethe decoder knows from the transmitted clipped prediction error the signof the clipping error. For this embodiment, the sign of the clippingerror is positive when the transmitted clipped prediction is +30 (or +31in case of an unreduced clip range), and negative when the transmittedclipped prediction is −31 (or −32 in case of an unreduced clip range).

[0125] A preferred PCM-bit substitution is as follows. If the unclippedprediction error e_(enc) is in the range [+30 . . . +37], the valuee_(enc)−30 is transmitted in PCM bits; in case of an unreduced cliprange, we have respectively the range [+31 . . . +38] and thetransmitted value e_(enc)−31. If e_(enc) is in the range [−38 . . .−31], the value +31−e_(enc) is transmitted as PCM-bits; in case of anunreduced clip range, we have respectively the range [−39 . . . −32] andthe transmitted value 32−e_(enc). If the prediction error e_(enc) islarger than +37 or smaller than −38 (larger than +38 or smaller than −39for an unreduced clip range), a value of 7 is transmitted in thePCM-bits.

[0126] This is a clipping operation that clips the absolute clippingerror to the range [0 . . . 7]. The clipping of the prediction errors tothe range [−31 . . . +30] in this embodiment (or [−32 . . . +31] for anunreduced clip range) is executed in a first clipper, whereas theclipping of the absolute clipping error to the range [0 . . . 7] isexecuted in a second clipper.

[0127] When the decoder at the second location SL receives a clippedprediction error of +30 (+31 for an unreduced clip range), it adds tothe output of the DPCM decoder loop the value that is transmitted in thePCM-bits. For a received/clipped prediction error of −31 (−32 for anunreduced clip range), the decoder subtracts from the output of the DPCMdecoder loop the value which is transmitted in the PCM-bits.

[0128] The output of the DPCM decoder loop is {tilde over(x)}_(dec)+(e_(dec))_(c). The value that is transmitted in the PCM-bitsin the case that PCM-bit substitution effectively took place isMAX{|e−(e_(enc))_(c)|,7} in this embodiment with 3 PCM-bits. When theabsolute clipping error is less than 8, the output of the decoder isthen

{circumflex over (x)} _(dec)+(e _(enc))_(c)+(e−(e ^(enc))_(c))

[0129] When there have been no transmission errors, the output is then{circumflex over (x)}_(enc)+e, which is exactly the value of theoriginal sample of the first digitized television IF signal.

[0130] When the PCM-bits have been used for the transmission of theabsolute clipping error (clipped in its turn to [0 . . . 7] in thisembodiment), the decoder at the second location SL does not know whatshould be the value for the LSB's when LSB's and MSB's are combined toan output sample in the combiner. Therefore, at the second location SL,a value of preference 4 is the three-bit LSB-input of the combiner CBfor a sample for which a PCM-bit substitution effectively took place.This reduces the average error in the three LSB's or PCM-bits of asample with PCM-bit substitution, because they have approximately auniform distribution in the range [0 . . . 7]. The output value of 4 isan arbitrary choice; a value of 3 is a good alternative.

[0131]FIG. 11 shows the implementation of PCM-bit substitution. A firstlocation PCM-bit substitution control signal SC1 controls thesubstitution of PCM-bits. The control signal SC1 is derived in a firstlocation clipping detector BSC1 from the clipped prediction error(e_(enc))_(c). The control signal SC1 is only true for (e_(enc))_(c)equal to +30 or −31 (or +31, respectively −32 for an unreduced cliprange). In substitutor BS1, the LSB's are replaced by a substitutionvalue CE emanating from the prediction error clipper PEC1 if the controlsignal SC1 is true.

[0132]FIG. 22 is a block diagram of the prediction error clipper PEC1. Afirst clipper CLP1 clips the prediction error e_(enc). A subtractorSUB13 calculates the clipping error and an absolute value calculator ABStakes the absolute value from it. A second clipper CLP2 clips theabsolute clipping error to the range [0 . . . 2(M−N)−1], generating thesubstitution value CE, being a clipped absolute clipping error.

[0133] A second location PCM-bit substitution control signal SC2controls an MSB corrector COR and a second location PCM-bits switch BS2.The control signal SC2 and a sign signal SGN are derived in a clippingdetector BSC2 from the clipped prediction error (e_(dec))_(c). Thecontrol signal SC2 is true for (e_(dec))_(c) equal to +30 or −31 (+31 or−32 for an unreduced clip range). The sign signal SGN indicates apositive clipping error if (e_(dec))_(c) is +30 and a negative clippingerror if (e_(dec))_(c) is −31 (+31, respectively −32 for an unreducedclip range).

[0134] A second location PCM-bits switch BS2 switches the M−N LSB's of asample of the second television IF signal between the received M−N bitsfrom the residual bit stream RTS and a fixed replacement (M−N)-bit word2^((N−M−1)), or 4 in the preferred embodiment. If control signal SC2indicates that PCM-bit substitution has taken place and if sign signalSGN indicates a positive clipping error, the MSB corrector COR2 adds theabsolute clipping error received via the residual bit stream RTS to theoutput {tilde over (x)}_(dec) of the DPCM decoder core D2. Otherwise,for sign signal SGN indicating a negative clipping error in case ofPCM-bit “substitution” the MSB corrector COR2 subtracts the absoluteclipping error received via RTS from {tilde over (x)}_(dec). The outputof the MSB corrector COR2 is applied to the combiner CB.

[0135] Instead of the implementation of FIG. 11, the correction can bein the loop LP21 of the DPCM decoder, and, consequently, also in theloop LP12 of the local decoder in the encoder. An implementation withthe correction of the PCM-bit substitution is given in FIG. 23. Theoutput of (COR1, COR2) is fed back to the predictor (PR1, PR2) In theencoder, BSC1 is replaced by BSC12, which is exactly the same clippingdetector BSC2 as in the decoder. The second improving part, namedPCM-bit substitution, has been described above referring to FIG. 11corresponding to the case whereby DPCM is implemented. FIG. 12 shows ablock diagram, illustrating a partial implementation of the firstembodiment of this invention, whereby the first and second improvementsare combined, HDPCM being implemented for transport of the N mostsignificant bits and PCM-bit substitution being carried out on the M−Nleast significant bits.

[0136] Instead of a clipped absolute clipped error, some othersubstitution value (CE) which is a function of the clipping error can beused for substitution of the PCM-bits, as long as that substitutionvalue (CE) can be represented by M−N bits. This function comes in theplace of the second clipper CLP2. In this case, the decoder has to mapby means of a second function the values received as residual transportsample to a value that can be added to or subtracted from the output ofthe DPCM decoder loop, if the control signal SC2 indicates that PCM-bitsubstitution has taken place. This mapping by means of a second functionhas to take place in the corrector COR2.

[0137] Instead of substituting all (M−N) PCM-bits, it is also possibleto substitute only P least significant bits of the (M−N) PCM-bits, withP smaller than (M−N). The other (M−N−P) bits can be used to transmit theM−N−P most significant bits of the (M−N) PCM-bits. In this case, in thesecond clipper CLP2, the absolute clip error has to be clipped to arange [0 . . . (2^(P)−1)], or a value which is a function of theabsolute clip error and which can be represented by P bits has to becalculated. In BS1 only P of the M−N PCM-bits are substituted if SC1indicates so, and SC1 is generated by BSC1 or BSC12 as described before.In the case that PCM-bit substitution has taken place, COR1 and COR2 addor subtract the value transmitted in those P bits (if necessary derivedby means of a second function) to its input, and BS2 replaces then theseP bits by the value 2^((P−1)) (or 2^((P−1))−1).

[0138] Third Improving Part, Named “Dynamic Clip Range Shifting”

[0139] Even with PCM-bit substitution, big clipping errors can stillhappen at sudden changes of the sampled television IF signal. Thesecoding errors can lead to visible or audible distortions. FIG. 13 andFIG. 14 illustrate such sudden changes of a television IF signal (markedIF). The vertical axis IFL (IF signal level) represents the amplitude ofthe television IF signal, and the horizontal axis (marked t) representsa time axis. FIG. 13 illustrates a television IF signal with negativemodulation having at time instant T_(SC) a sudden change (or transition)from a low amplitude (a bright image part) to a high amplitude (a darkimage part). FIG. 14 illustrates a television IF signal with negativemodulation having at time instant T_(SC) a sudden change (or transition)from a high amplitude (a dark image part) to a low amplitude (a brightimage part). Both FIG. 13 and FIG. 14 show samples S_(n) of thetelevision IF signal each corresponding to a time instant T_(n). S_(i),at T_(i), is the sample taken just before, and S_(i+1), at T_(i+1), thefirst sample taken just after the sudden change at time instant T_(SC).Graph IFE (with dashed lines) between T_(sc) and T_(i+1) represents theexpected television IF signal immediately after T_(sc). (S_(i+1))_(exp)represents the first sample after T_(sc) as it is expected. It is clearfrom FIG. 13 and FIG. 14 that there is a relatively big prediction error(or e) at T_(i+1) between the sample S_(i+1) (or X) and the expected (orpredicted) sample amplitude (S_(i+1))_(exp) (or {circumflex over (x)}).

[0140] To avoid big clipping errors caused by sudden changes of thesampled television IF signal, an additional improvement is obtained bymaking the prediction error clip range not fixed but adaptive, applyinga method called here dynamic clip range shifting. The said dynamic cliprange shifting is related to the phase of the IF carrier and to theluminance of the television IF signal. Therefore, the IF carrier phaseand the luminance are estimated in what is further called respectivelyan IF carrier estimation and a luminance estimation.

[0141]FIG. 15 shows the dynamic clip range shifting added to theelementary set-up, consisting of an IF carrier estimation being aphase-locked loop (PLL1, PLL2), a luminance estimator (LUE1, LUE2) and ashift estimator (SHE1, SHE2) in both the first location FL and thesecond location SL.

[0142] First Divisional Part of Dynamic Clip Range Shifting, Called “IfCarrier Estimation”

[0143] The dynamic clip range shifting is based on the IF carrier phase.As the IF carrier phase is unknown in the decoder, the decoder has toestimate the IF carrier phase. In order to have the same shift inencoding and decoding with a dynamic clip range shifting, the encoderhas to do the same IF carrier estimation as the decoder. The IF carrierestimation is by preference realized by means of a phase-locked loopPLL1, PLL2, one in the encoder and one in the decoder. In both theencoder and the decoder, it is a decod d television IF signal {tildeover (x)}, respectively being the output {tilde over (x)}_(enc) of thelocal decoder in the encoder and the output {tilde over (x)}_(dec) ofthe decoder, which is the reference input of the PLL.

[0144] The PLL should estimate the phase of the IF carrier present inthe decoded television IF signal. In this embodiment, the IF carrier of45.75 MHz in the spectrum of the digitized NTSC M television signal ismirrored to a frequency of (3*16.2)−45.75=2.85 MHz because of the 16.2MHz sampling. For other transmission standards (e.g., PAL B/G with an IFcarrier frequency of 38.9 MHz), the mirrored IF carrier will have adifferent frequency (e.g., 6.5 MHz for PAL B/G).

[0145] In order to reduce the complexity, by preference a simple PLLstructure as illustrated in FIG. 16 is implemented, with which thezero-crossings of a signal, further referred to as the generated sinewave sin(Φ) and being generated by a sine wave generator SIN(Φ), arematched to the zero-crossings of the band-pass filtered decoded IFsignal {tilde over (x)}_(BP). The PLL phase Φ is a linear function oftime t, expressed as Φ=ω_(IF)t with ω_(IF) being the angular frequencyof the IF carrier. By “zero-crossings” are meant points of time where azero-mean signal passes from a positive to a negative value orvice-versa.

[0146] Because zero-crossings are time-critical for the good operationof the PLL, the decoded IF signal {tilde over (x)} is first band-passfiltered with a band-pass filter BPF before being applied to azero-crossing comparator ZCC in order to reduce the influence of theaudio signal. On FIG. 4, it is visible in the samples 237280 to 237305that, for this embodiment for the NTSC M transmission standard, theaudio signal, which is in the spectrum centered around 2.85 MHz+4.5MHz=7.35 MHz, resembles, referred to the 2.85 MHz IF carrier signal, arelatively noisy component which influences significantly thezero-crossings. A simple symmetrical three-taps low-pass filter withtaps 1/2/1 is sufficient as band-pass filter and is therefore preferred.Such a filter not only suppresses the audio signal around 2.85 MHz+4.5MHz=7.35 MHz in the digitized IF signal spectrum between 0 and 8.1 MHz,but also the color information around 2.85 MHz+3.58 MHz=6.43 MHz. In thecase of PAL B/G, this low-pass filter has to be replaced by a high-passfilter.

[0147] The precision of the phase values in the PLL of this embodimentis by preference 8 bit, which means that the phase range of [0°. . .360°[ is to be mapped on the range [0 . . . 256[. For NTSC M, a 2.85 MHzsine wave sampled at 16.2 MHz, an increment between two consecutivesamples corresponds to a phase increment ΔΦ of (256*2.85)/16.2 orapproximately 45. For the phase represented by 8 bits, the fixed phaseincrement is ΔΦ=45 for NTSC M (it is 103 for PAL B/G). The phase valuesare kept within the range [0 . . . 255] through a modulo-256 operation,which in a 8-bit-adder happens automatically.

[0148] The PLL as shown in FIG. 16 is a loop which consists of aregister REG1, an adder ADDP which adds every clock cycle a value ΔΦ tothe PLL phase Φ, and extra logic to align the zero-crossings of thehigh-pass filtered decoded IF signal {tilde over (x)}_(BP) and thelocally generated sine wave sin(Φ). This extra logic includes thezero-crossing comparator ZCC and a Φ adjustment ΦADJ.

[0149] The sign bit of a sine value gives information on thezero-crossings of that sine wave. If, for a given sample, the sign ofthe band-pass filtered decoded IF signal {tilde over (x)}_(BP) is equalto the sign of the generated sine wave sin(Φ), the Φ adjustment ΦADJpasses its input Φunchanged to its output.

[0150] If, for a given sample, the sign of the band-pass filtereddecoded IF-signal {tilde over (x)}_(BP) is not equal to the sign of thegenerated sine wave sin(Φ), the PLL phase Φ is adjusted in ΦADJ by theminimum possible amount such that the sign of the band-pass filtereddecoded IF-signal {tilde over (x)}_(BP) and the sign of the generatedsine wave sin(Φ) become the same as listed in Table 1. TABLE 1 Band-passfiltered decoded IF signal sine wave PLL phase Φ {tilde over (x)}BP sin(Φ) PLL phase Φ adjusted to positive negative [128 . . . 191] 127positive negative [192 . . . 255] 256 modulo 256 = 0 negative positive [0 . . . 63] (−1) modulo 256 = 255 negative positive  [64 . . . 27] 128

[0151] The phase adjustment as described here leads to fast recovery ofthe exact PLL phase Φ in the decoder at start-up or after a transmissionerror.

[0152] In this embodiment, the output of the zero-cross comparator ZCCindicates whether the signs of {tilde over (x)}_(BP) and sin(Φ) areidentical or not. The Φ-adjustment ΦADJ adjusts the PLL phase Φaccording to Table 1, or passes Φ unchanged from input to output if{tilde over (x)}_(BP) and sin(Φ) have the same sign.

[0153] The band-pass filter BPF introduces a phase shift between itsoutput and its input and consequently also a phase shift to the outputof the PLL. This phase shift by the band-pass filter BPF must becompensated when the PLL phase Φ is used for decisions on the shift ofthe clip range. With the here preferred three-tap band-pass filter, theshift introduced by the band-pass filter is one cycle of a sample clock,or ΔΦ units of the 8-bit-precision phase.

[0154] Second Divisional Part of Dynamic Clip Range Shifting, Called“Luminance Estimation”

[0155] As said before, the dynamic clip range shifting is not onlyrelated to the phase of the IF carrier, but also to the luminance of thecomposite signal that modulates the IF carrier. The luminance is howevernot transmitted as side information within the sampled televisionIF-signal and must therefore be estimated in both the encoder and thedecoder. In order to have the same shift in encoding and decoding with adynamic clip range, the encoder must have the same luminance estimationas the decoder.

[0156] A television IF signal is approximately an IF carrier which ismodulated in amplitude by the luminance, and is represented here furtheras Y*sin(ω_(IF)t) whereby Y is the luminance in the case of positivemodulation or the inverted luminance in the case of negative modulation,and whereby ω_(IF) is the angular frequency of the IF carrier. Aftersampling at 16.2 MHz, the spectrum of the modulated IF carrier ismirrored to the range [−8.1 . . . +8.1 MHz].

[0157] The luminance estimation is by preference carried out in twosteps as described below.

[0158] The first step of the luminance estimation is the multiplicationof {tilde over (x)}_(BP2), the band-pass filtered output signal x of thedecoder or of the local decoder in the encoder, by an estimated IFcarrier. The estimated IF carrier is derived from the PLL phase Φ, and{tilde over (x)}_(BP2) is derived from {tilde over (x)} by means of theband-pass filter BP2.

[0159] The output Φ of the PLL has a phase delay of one sample periodwith respect to the decoded IF signal x because of the delay caused bythe three-taps band-pass filter BPF in the PLL. The phase delay of onesample period can be compensated for in the luminance estimation by notmultiplying sin(Φ) but sin(Φ+ΔΦ) with {tilde over (x)}_(BP2). However,the band-pass filter BPF2 causes also a delay in the other input {tildeover (x)}_(BP2) of the mixer. In general, sin(Φ+n*ΔΦ) with n thedifference of delay in sample units between both inputs of the mixer, isan estimation for the IF carrier sin (ω_(IF)*t). The multiplication isthen

{tilde over (x)}*sin (ω _(IF) *t) Y*sin(ω_(IF) *t)*sin(ω_(IF)*t)=0.5*Y*(1−cos(2ω_(IF) *t))

[0160] The second step of the luminance estimation is a low-passfiltering by means of a low-pass filter LPF of the result of themultiplication of the first step of the luminance estimation in order tosuppress the high-frequency component cos(2ω_(IF)*t) and to obtain the(inverted) luminance Y, being an estimated (inverted) luminance L.

[0161] The output of the mixing of the decoded IF signal x and the sinewave sin(Φ+ΔΦ) has a strong component at the double IF carrierfrequency. With this embodiment for NTSC M, the said double IF carrierfrequency is 2.85*2=5.7 MHz. The said low-pass filtering LPF of thesecond step of the luminance estimation is by preference a 5-tap filterwith taps 1/1/2/1/1 for NTSC M. This filter without multipliers ispreferred because of its simple realization, and because it almostcompletely suppresses the unwanted mixing output of 5.7 MHz.

[0162]FIG. 17 shows a block diagram of the luminance estimation part(marked LUE) of this embodiment of the invention as described above,with which the estimated (inverted) luminance L is obtained from the PLLphase Φ and the decoded IF signal {tilde over (x)}_(BP2). The saidmultiplication of the first step of the luminance estimation is carriedout as a mixing in a mixer MIX of the band-pass filtered decoded IFsignal {tilde over (x)}_(BP2) with a local oscillator sine wavesin(Φ+n*ΔΦ) from a local oscillator SIN(Φ+v*ΔΦ). The output of low-passfilter LPF is the estimated (inverted) luminance L.

[0163] The band-pass filter BPF2 is in this embodiment by preference thesame filter as BPF. In this preferred embodiment, {tilde over (x)}_(BP2)is equal to {tilde over (x)}_(BP)PLL.

[0164] The band-pass filter BPF2 suppresses the audio and chrominancecomponents because they lead to mixer products that are not sufficientlysuppressed by the low-pass filter LPF.

[0165] Third Divisional Part of Dynamic Clip Range Shifting Is DescribedNow, Named “Shift Calculation”

[0166] A preferred calculation of a shift of the clip range isdescribed, based on the IF carrier phase and luminance estimationdescribed above.

[0167] Prediction errors can lie outside the clip range at unpredictablesudden changes or transitions of the television IF signal, of which somehave been described before with the aid of FIG. 13 and FIG. 14. In thesampled television IF signal, these said sudden changes are maximal whena difficult-to-predict sample coincides with a peak of the sine wave ofthe IF carrier. Therefore, a shift of the clip range ([−31 . . . +30]for this embodiment with reduced clip range) must be in proportion tothe instantaneous value of the sine wave of the IF carrier which isrepresented as sin(ω)_(IF)*t).

[0168] In order to avoid a multiplication in the calculation of theshift sh of the clip range as represented by

sh(t)=C*sin(ω_(IF) *t)

[0169] whereby C is an arbitrary constant, the estimated IF carrier isquantized. A preferred implementation uses seven levels of quantization.

[0170] The preferred thresholds for the 7-level quantization are givenin the first column of Table 2. TABLE 2 Instantaneous value shift amountshift amount of the IF carrier at low at high sin(ω_(IF) * t) luminanceluminance    256 * sin(ω_(IF) * t) ≦ −96 +12 −12 −96 < 256 *sin(ω_(IF) * t) ≦ −80 +8 −8 −80 < 256 * sin(ω_(IF) * t) ≦ −64 +4 −4 −64< 256 * sin(ω_(IF) * t) < 64 0 0   64 ≦ 256 * sin(ω_(IF) * t) < 80 −4 +4  80 ≦ 256 * sin(ω_(IF) * t) < 96 −8 +8   96 ≦ 256 * sin(ω_(IF) * t) −12+12

[0171] The amount of shift of the prediction error clip range is bypreference determined as follows for the case of negative modulation.

[0172] When the luminance is low, a hardly predictable transition is atransition to a high luminance. When the first sample of the IF signalafter such a transition from a low to a high luminance coincides with apositive peak of the IF carrier as illustrated in FIG. 14, theprediction error will have a large negative value, for some transitionsof this embodiment much less than −31. Therefore, the losslessness ofthe compression in case of hardly predictable luminance transitions isimproved by introducing a shift sh of the before mentioned clip range of[−31 . . . +30] for samples corresponding to a dark luminance to a cliprange of [(−31+sh) . . . (30+sh)] whereby sh is a negative value of bypreference 0, −4, −8 or −12, corresponding respectively to the beforementioned quantization levels of the instantaneous value of the IFcarrier as indicated in Table 2. When the first sample of the IF signalafter such a transition from a low to a high luminance coincides with anegative peak of the IF carrier, the shift should have a positive value,for this embodiment by preference with sh values of 0, +4, +8 and +12corresponding to the quantization levels of the quantized instantaneousvalue of the estimated IF carrier as indicated in Table 2. In FIG. 9,SCL9 illustrates a shift of the clip range [−32 . . . +31] for a shiftsh of +8.

[0173] When the luminance is high, a hardly predictable transition is atransition to a low luminance. Similarly as with the above describedtransition from a low to a high luminance, coding errors are reduced bya shift sh of the clip range. When the first sample of the IF signalafter such a transition from a high to a low luminance coincides with apositive peak of the IF carrier as illustrated in FIG. 13, a positiveshift sh of the clip range is needed. When the first sample of the IFsignal after a transition from a high to a low luminance coincides witha negative peak of the IF carrier, a negative shift sh of the clip rangeis needed. Table 2 shows the preferred shift amounts for this embodimentin function of the quantized instantaneous value of the estimated IFcarrier.

[0174] The shift values of Table 2 are for the case of negativemodulation, as is the case for NTSC M. For negative modulation, a lowluminance corresponds to high values of L, and a high luminancecorresponds to low values of L. For positive modulation, the signs ofthe shift amount should be inverted, because for positive modulation, alow luminance corresponds to low values of L, and a high luminancecorresponds to high values of L.

[0175] Improvement to the Shift Calculation, Named “Dejitter”

[0176] Practice has shown that it is preferred to add some dejitter tothe thresholds of the low and high luminance values of the estimatedluminance L. With this embodiment, the dejitter is obtained by takinginto account both the estimated (inverted) luminance of the actualsample and the estimated luminance of the previous sample. A sample isconsidered to belong to a high luminance area if the estimated invertedluminance L of that sample is less than 8 and the estimated luminance Lof the previous sample is less than 8−2 or 6. Similarly, a sample isdetermined to belong to a low luminance area if the estimated invertedluminance L of that sample is higher than 12 and the estimated luminanceL of the previous sample is higher than 12+2 or 14. The just mentionedthresholds of low and high luminance are given here as an example butdepend on the reference voltage used for the analog-to-digitalconversion of the television IF signal. The estimated luminances changein accordance with a change of the reference voltage. The thresholdsshould be exchanged if the modulation is positive instead of negative.The shift of the clip range for a given sample depends on the level ofthe quantized IF carrier. That quantized IF carrier depends on the PLLphase Φ from the phase-locked loop PLL. The phase used to decide on theshift of the clip range contains compensation for a number of delays.These delays are illustrated in FIG. 20. FIG. 20 shows five graphs G1,G2, G3, G4 and G5. Graph G1 is used as time reference for the othergraphs of the same figure and shows a block signal with a period(sampling period T_(B)) and phase corresponding to the sampling rate ofthe samples x. Graph G2 represents the consecutive incoming samplesx_(i), x_(i+1), Graphs G3, G4 and G5 illustrate the following delays:

[0177] a two-sample delay between the shift of the clip range of theprediction errors and the DPCM output (this delay depends on theimplementation of the DPCM loops), illustrated in graph G3;

[0178] a one-sample delay caused by the three-tap band-pass filter BPFin the PLL, illustrated in graph G4;

[0179] a one-sample delay between the calculation of the shift amountand the actual shifting of the clip range, illustrated in graph G5 (thisdelay too depends on the implementation of the embodiment).

[0180] For the considered implementation, there is a compensation for atotal delay of four samples. Therefore, the sine wave used for thedecision on the shift of the clip range is for this embodiment expressedby sin(Φ+4ΔΦ) or, for NTSC M, sin (Φ+4*(256*2.85/16.2)), which issin(Φ+180) for an B-bit representation of the phase.

[0181]FIG. 18 shows a block diagram of the shift estimator (marked SHE)of this embodiment of the invention as described above, with which theshift sh is obtained from the PLL phase Φ and the estimated (inverted)luminance L. From the PLL phase Φ, a sine wave sin(Φ+4ΔΦ) is generatedwith a sine wave generator SIN(Φ+4ΔΦ). The sine wave sin(Φ+4ΔΦ) isquantized in a 7-level quantizer Q of which the output is applied to ashift calculator SHC. In the shift calculator SHC, the amount of shiftsh of the prediction error clip range is determined as described above.

[0182] The implementation of the dejitter in this embodiment includes aregister REG2 which delays the estimated luminance L of a sample for theduration of one sample.

[0183] The dynamic clip range shifting is by preference implemented inthree consecutive steps.

[0184] subtract the shift from the prediction error, what results in ashifted prediction error;

[0185] clip the shifted prediction error to a fixed clip range, for thisembodiment to the range [−31 . . . +30] (or [−32 . . . +31] in the caseof unreduced clip range), what results in a clipped shifted predictionerror;

[0186] add the shift amount to the clipped shifted prediction error.

[0187]FIG. 19 illustrates the implementation of the dynamic clip rangeshifting. At the first location, a subtractor SUB12 subtracts the shiftsh_(enc) from the prediction error e_(enc), resulting in a shiftedprediction error e_(enc)−sh_(enc). The shifted prediction errore_(enc)−sh_(enc) is clipped in the prediction error clipper PEC1 andresults in a clipped shifted prediction error (e_(enc)−sh_(enc))_(c).The shift sh_(enc) is added by adder ADD13 to the clipped shiftedprediction error (e_(enc)−sh_(enc))_(c) and the resulting sum(e_(enc))_(c) is added by adder ADD11 to the encoder prediction{circumflex over (enc)} to result in the output {tilde over (x)}_(enc)of the local decoder in the encoder. At the second location, the shiftsh_(dec) is added in an adder ADD22 to the shifted decoder output {tildeover (x)}_(dec)−sh_(dec) resulting in the decoder output {tilde over(x)}_(dec).

[0188] HDPCM is not implemented in FIG. 19; it is however possible tocombine the dynamic clip range shifting with HDPCM as described beforebecause the range of transmitted prediction errors remains [−31 . . .+30]. FIG. 5, which is a complete implementation of this embodiment,shows HDPCM combined with dynamic clip range shifting.

[0189] Extra-Improving Part to the Combination of Dynamic Clip RangeShifting and HDPCM

[0190] The implementation of FIG. 19 suggests that the shift sh can beconsidered as an improvement of the predictor {tilde over (x)}. As aconsequence, the input to the prediction mapper (PM1, PM2) of HDPCM whenHDPCM and dynamic clip range shifting are combined can be ({tilde over(x)}+sh) instead of {tilde over (x)}. In this way, not only differencesin prediction {tilde over (x)} but also differences in shift sh betweenencoder and decoder are also corrected by HDPCM.

[0191] The above described first embodiment of the present invention canbe implemented in a programmable logic device, by preference working ata clock frequency equal to the sampling frequency of the first digitizedtelevision IF frequency.

[0192] Although the present invention has been described with referenceto the first embodiment of the present invention, the invention is notlimited solely to this first embodiment. Digitized television IF signalsaccording to the same or other transmission standards than the onementioned in the description of the first embodiment, may be transportedover the same or other transport channels than the one mentioned in thedescription, the prediction and the dynamic clip range shifting beingadapted to the transmission standard. Another digital signal than adigitized television IF signal can also be transported, again with theprediction and the dynamic clip range shifting being adapted to theparameters of the digital signal. Other values can be chosen for M, Nand C (or even P). Moreover, various other preferred choices in theimplementation of the first embodiment of the present invention can bemade differently.

[0193] The first embodiment of the present invention has been describedabove starting from the implementation of an elementary implementationwith reference to FIG. 6, and followed by a step-by-step addition ofimproving parts to a full implementation shown in FIG. 5. By theimproving parts is thereby meant HDPCM, PCM-bit substitution and dynamicclip range shifting.

[0194] The invention can, at the expense of a lower grade transport ofthe digital signal or at the expense of a higher probability for codingerrors, be implemented as a combination of the elementary implementationas described in the description of the first embodiment and one or morebut not all of the improving parts described in the description of thefirst embodiment. The below described second and third embodiments ofthe present invention are two examples of combinations of the elementaryimplementation and one or more of the improving parts.

[0195] Second Preferred Embodiment

[0196] The second embodiment of the present invention is a method forthe transport of a sample S1 of a first digital signal over a transportchannel TC from a first location FL to a second location SL, where it isreceived as a sample S2 of a second digital signal. The to betransported digital signal and transport channel TC are in particular(but not necessarily) the same as described in the first embodiment, theto be transported digital signal thus being a digitized television IFsignal.

[0197] The second embodiment is described below with reference to andcompared to the description of the first embodiment of this inventionand the figures used for it.

[0198]FIG. 21 shows a block diagram of the implementation of the secondembodiment. The second embodiment is characterized in that the M−N LSB'sare transmitted uncompressed, by preference in PCM, and that the N MSB'sare compressed to N−C bits by implementation of the improving part HDPCMwith the HDPCM extra-improving part “reduction of the clip range” and/orthe HDPCM extra improving part “non-uniform-mapping”. In the blockdiagram of FIG. 21 is thus combined the elementary set-up of FIG. 6 withHDPCM as illustrated in FIG. 10; the extra-improving parts are notvisible in FIG. 21.

[0199] For a more detailed description of this second embodimentreference is made to the description of the first embodiment, whereinall the parts of this second embodiment are described.

[0200] Third Preferred Embodiment

[0201] The third embodiment of the present invention is a method for thetransport of a sample S1 of a first digital signal over a transportchannel TC from a first location FL to a second location SL, where it isreceived as a sample S2 of a second digital signal. The to betransported digital signal and transport channel TC are in particular(but not necessarily) the same as described in the first embodiment, theto be transported digital signal thus being a digitized television IFsignal.

[0202] The third embodiment is described below with reference to and incomparison to the description of the first embodiment of this inventionand the figures used for it.

[0203]FIG. 15, already referred to in the description of the firstembodiment, shows a block diagram of the implementation of the thirdembodiment. The third embodiment is characterized in that the M−N LSB'sare transmitted uncompressed, by preference in PCM, and that the N MSB'sare compressed to N−C bits by implementation of the improving part“dynamic clip range shifting”. In the block diagram of FIG. 15 is thuscombined the elementary set-up of FIG. 6 with dynamic clip rangeshifting as illustrated in FIG. 19.

[0204] For a more detailed description of this third embodimentreference is made to the description of the first embodiment, whereinall the parts of this third embodiment are described.

[0205] Presented after the following Reference Key are the claims, whichdefine the present invention exemplary embodiments of which arepresented above without limitation.

[0206] Reference Key

[0207] Reference Sign Designation of Indicated Feature

[0208] ABS absolute value calculator

[0209] AD analog-to-digital converter

[0210] ADD . . . adder . . .

[0211] ADDP adder in PLL

[0212] AS1 first analog signal

[0213] AS2 second analog signal

[0214] BS1 first location substitutor

[0215] BS2 second location substitutor

[0216] BSC1 first location clipping detector

[0217] BSC2 second location clipping detector

[0218] C clipped (index), also noted as ( . . . )_(c)

[0219] CB combiner

[0220] CE substitution value

[0221] CLP clipper

[0222] COR PCM-bit substitution MSB corrector

[0223] CTS compressed transport sample

[0224] D1 encoder DPCM-core

[0225] D2 decoder DPCM-core

[0226] DA digital-to-analog converter

[0227] dec of decoder (index)

[0228]1 encoder DPCM-core

[0229] D2 decoder DPCM-core

[0230] DA digital-to-analog converter

[0231] DS1 first digital signal

[0232] DS2 second digital signal

[0233] e prediction error

[0234] enc of encoder (index)

[0235] f frequency (axis)

[0236] FL first location

[0237] Φ PLL phase

[0238] ΦADJ Φ-adjustment

[0239] G1 . . . G5 graph 1 . . . graph 5

[0240] H(f) spectrum amplitude (axis)

[0241] BPF band-pass filter

[0242] IFL If signal level

[0243] L estimated luminance

[0244] LSB least significant bits (PCM-bits)

[0245] LP11 first loop of the encoder

[0246] LP12 second loop of the decoder

[0247] LP21 loop of the decoder

[0248] LUE luminance estimator

[0249] m( . . . ) mapped value of ( . . . )

[0250] MIX mixer

[0251] MSB most significant bits (DPCM-bits)

[0252] PEC prediction error clipper

[0253] PLL phase-locked loop

[0254] PM prediction mapper

[0255] PR predictor

[0256] Q quantizer

[0257] Q&CL quantizer and clipper

[0258] REC receiver

[0259] REG1 register (in PLL)

[0260] REG2 register (in shift estimator)

[0261] RTS residual transport sample

[0262] S1 sample of the first digital signal (sample of a digitizedtelevision IF signal in the embodiments)

[0263] S2 sample of the second digital signal (sample of a digitizedtelevision IF signal in the embodiments)

[0264] SC1 first location PCM-bit substitution control signal

[0265] SC2 second location PCM-bit substitution control signal

[0266] SCL . . . scale

[0267] sh shift (of the clip range)

[0268] SGN sign signal

[0269] SHC shift calculator

[0270] SHE shift estimator

[0271] S_(i) i-th sample of the television IF signal

[0272] SL second location

[0273] SN sample number (axis)

[0274] S_(n) n-th sample of the television IF signal

[0275] SP splitter

[0276] SUB subtractor

[0277] SV sample value (axis)

[0278] t time (axis)

[0279] TRA transmitter

[0280] TC transport channel

[0281] TCBS transport channel bit-stream

[0282] T_(S) sampling period

[0283] VLC variable length coder

[0284] VLD variable length decoder

[0285] wr ( . . . ) wrapped-around value of ( . . . )

[0286] x sample (of a digital signal to be transported)

[0287] {circumflex over (x)} prediction (of a decoder, or of a localdecoder in an encoder)

[0288] {tilde over (x)} output signal(of a decoder, or of a localdecoder in an encoder)

[0289] ZCC zero-cross comparator

What is claimed is:
 1. Method for the transmission of a first digitalsignal (DS1) from a first location (FL) over a transport channel (TC) toone or more second locations (SL) where it is received as a seconddigital signal (DS2) which is substantially equal to the first digitalsignal (DSl), whereby a sample (S1) of the first digital signal (DSl)represented by M bits, being the total of N most significant bits (MSB)and M−N least significant bits (LSB), is transported over the transportchannel (TC) as a transport sample comprising at least two parts, ofwhich one part is a compressed transport sample (CTS) represented by N−Cbits, N being smaller than M and C being smaller than N and larger than0, the N−C bits being obtained through predictive coding of the N mostsignificant bits of the sample (S1) of the first digital signal (DS1),whereby to each sample (S1) of the first digital signal (DS1)corresponds at least a prediction ({circumflex over (x)}_(enc))representing the predicted N most significant bits of the sample (S1) ofthe first digital signal (DS1) whereby the prediction is based onpreviously compressed samples, a prediction error (e_(enc)) representingthe difference between the N most significant bits of the sample (S1) ofthe first digital signal (DS1) and the said prediction ({circumflex over(x)}_(enc)), and a clipping error representing the difference betweenthe prediction error (e_(enc)) and a clipped prediction error((e_(enc))_(c)) which is the prediction error (e_(enc)) clipped by afirst clipper (PEC1) to a clip range represented as [A . . . B] by meansof N−C bits, A and B being integers and B−A being equal to or smallerthan 2^((N−C)) ⁻¹, and the other part is a residual transport sample(RTS) represented by M−N bits, characterized in that the said residualtransport sample (RTS) is either equal to the M−N least significant bits(LSB) of the said sample (S1) of the first digital signal (DS1) in thecase that the prediction error (e_(enc)) corresponding to the saidsample (S1) of the first digital signal (DS1) is in the range [A . . .B], or, in the other case, the P least significant bits of the M−N LSB'sare equal to a substitution value (CE) which is a quantisation functionof the absolute clipping error corresponding to the said sample (S1) ofthe first digital signal (DS1), whereby the number of output levels ofthe said quantisation function of the absolute clipping error is equalto or less than 2^(P), whereby P is equal to or less than M−N andwhereby the other M−N−P bits of the M−N bits of the RTS are equal to theM−N-P most significant bits of the M−N LSB's of the said sample (S1) ofthe first digital signal (DS1).
 2. Method according to claim 1, whereinthe compressed transport sample (CTS) is the clipped prediction error((e_(enc))_(c)).
 3. Method according to claim 1, wherein the compressedtransport sample (CTS) is an in the range [A . . . B] wrapped around sumof the clipped prediction error ((e_(enc))_(c)) and a mapped value(m({circumflex over (x)}_(enc))) of the prediction ({circumflex over(x)}_(enc)), and that the said mapped value (m({circumflex over(x)}_(enc))) of the prediction ({circumflex over (x)}_(enc)) is theprediction ({circumflex over (x)}_(enc)) mapped on a range [D . . . E],E and D being integers and E−D being equal to or smaller than 2^((N−C))⁻¹.
 4. Method according to claim 3, wherein the prediction ({circumflexover (x)}_(enc)) is mapped or quantized in a non-uniform way, such thatthe quantization is fine for prediction values corresponding to smallinput amplitudes and rough for prediction values corresponding to bigamplitudes of the first digital signal.
 5. Method for the transmissionof a first digital signal (DS1) from a first location (FL) over atransport channel (TC) to one or more second locations (SL) where it isreceived as a second digital signal (DS2) which is substantially equalto the first digital signal (DS1), whereby a sample (S1) of the firstdigital signal (DS1) represented by N bits, is transported over thetransport channel (TC) as a transport sample represented by N−C bits, Cbeing smaller than N and larger than 0, the N−C bits being obtainedthrough predictive coding of the N bits of the sample (S1) of firstdigital signal (DS1), whereby with each sample of the first digitalsignal (DS1) corresponds at least a prediction ({circumflex over(x)}_(enc)) representing the predicted N bits of the sample (S1) of thefirst digital signal (DS1), whereby the prediction is based onpreviously compressed samples, and a prediction error (e_(enc))representing the difference between the N bits of the sample (S1) of thefirst digital signal (DS1) and the said prediction ({circumflex over(x)}_(enc)), characterized in that the transport sample is an in therange [A . . . B] wrapped around sum of the prediction error (e_(enc))clipped to a fixed range named clip range [A . . . B], A and B beingintegers and B-A being equal to or smaller than 2^((N−C)) ⁻¹, and amapped value (m(x enc)) of the prediction ({circumflex over (x)}_(enc))which has been mapped on a range [D . . . E], E and D being integers andE−D being equal to or smaller than 2^((N−C)) ⁻¹.
 6. Method according toclaim 5, wherein the prediction (x enc) is mapped or quantized in anon-uniform way, such that the quantization is fine for predictionvalues corresponding to small input amplitudes and rough for predictionvalues corresponding to big amplitudes of the first digital signal. 7.Method for the transmission of a first digital signal (DS1) from a firstlocation (FL) over a transport channel (TC) to one or more secondlocations (SL) where it is received as a second digital signal (DS2)which is substantially equal to the first digital signal (DS1), wherebya sample (S1) of the first digital signal (DS1), represented by N bits,is transported over the transport channel (TC) as a transport samplerepresented by N−C bits, C being smaller than N and larger than 0, theN−C bits being obtained through predictive coding of the N bits of thesample (S1) of the first digital signal (DS1), whereby with each sample(S1) of the first digital signal (DS1) corresponds at least a prediction({circumflex over (x)}_(enc)) representing the predicted N bits of thesample (S1) of the first digital signal (DS1), whereby the prediction({circumflex over (x)}_(enc)) is based on previously compressed samples,and a prediction error (e_(enc)) representing the difference between theN bits of the sample (S1) of the first digital signal (DS1) and the saidprediction enc and whereby the transport sample is the prediction error(e_(enc)) clipped to a range named clip range, which can be representedby means of N−C bits, wherein the clip range is shiftable in the saidsample in function of one or more actual parameters of the said samplein the first digital signal.
 8. Method according to claim 7, whereby thefirst digital signal (DS1) is a digitized television IF signal with anIF carrier and featuring as parameters at least an estimated luminanceand an estimated IF carrier phase, the second digital signal (DS2)consequently also being a digitized television IF signal, wherein: theclip range [A . . . B] is shifted over a shift (sh) to [A+sh . . .B+sh], whereby the shift (sh) for a sample is determined by theestimated luminance and/or by the estimated IF carrier phase in thatsample, the absolute amount of the shift (sh) is proportional to theabsolute value of the amplitude of the estimated IF carrier, for atelevision IF signal with negative modulation, either the shift (sh) isnegative in case of a positive peak of the estimated IF carrier and alow luminance, and in case of a negative peak of the estimated IFcarrier and a high luminance, or the shift (sh) is positive in case of anegative peak of the estimated IF carrier and a low luminance, and incase of a positive peak of the estimated IF carrier and a highluminance, for a television IF signal with positive modulation, eitherthe shift (sh) is positive in case of a positive peak of the estimatedIF carrier and a low luminance, and in case of a negative peak of theestimated IF carrier and a high luminance, or the shift (sh) is negativein case of a negative peak of the estimated IF carrier and a lowluminance, and in case of a positive peak of the estimated IF carrierand a high luminance.
 9. Method for making the same estimation of the IFcarrier of a sampled IF signal in a receiver comprising a decoder and ina transmitter comprising an encoder which includes a local decoder,characterized in that in both the transmitter and the receiver, theoutput of the decoder or the output of the local decoder in the encoderis first band-pass filtered and then, the output of the band-pass filteris the input of a phase-locked loop which tracks the phase of the IFcarrier.
 10. Method for making the same estimation of the luminance ofan IF signal in a receiver comprising a decoder and in a transmittercomprising an encoder which includes a local decoder, characterized inthat first the IF carrier is estimated at both encoder and decoder, thatthen in both the transmitter and receiver the estimated IF carrier ismultiplied with the band-pass filtered output of the local decoder inthe encoder or with the band-pass filtered output of the decoder, andthat finally the multiplier result is low-pass filtered in bothtransmitter and receiver.
 11. Method according to claim 8, wherein thecompressed transport sample (CTS) is an in the-range [A . . . B] wrappedaround sum of the clipped prediction error ((e_(enc))_(c)) and a mappedvalue m(y) with y being either the prediction {circumflex over(x)}_(enc) or the sum of the prediction {circumflex over (x)}_(enc) andthe shift sh, and that the said mapped value m(y) is in a range [D . . .E], E and D being integers and E−D being equal to or smaller than2^((N−C)) ⁻¹, the mapping being either a uniform or a non-uniformmapping, the non-uniform mapping being such that the quantization isfine for prediction values corresponding to small input amplitudes andrough for prediction values corresponding to big amplitudes of the firstdigital signal.
 12. Method according to claim 1, whereby the firstdigital signal (DS1) is a digitized television IF signal with an IFcarrier and featuring as parameters at least an estimated luminance andan estimated IF carrier phase, the second digital signal (DS2)consequently also being a digitized television IF signal, wherein: theclip range [A . . . B] is shifted over a shift (sh) to [A+sh . . .B+sh], whereby the shift (sh) for a sample is determined by theestimated luminance and/or by the estimated IF carrier phase in thatsample, the absolute amount of the shift (sh) is proportional to theabsolute value of the amplitude of the estimated IF carrier, for atelevision IF signal with negative modulation, either the shift (sh) isnegative in case of a positive peak of the estimated IF carrier and alow luminance, and in case of a negative peak of the estimated IFcarrier and a high luminance, or the shift (sh) is positive in case of anegative peak of the estimated IF carrier and a low luminance, and incase of a positive peak of the estimated IF carrier and a highluminance, for a television IF signal with positive modulation, eitherthe shift (sh) is positive in case of a positive peak of the estimatedIF carrier and a low luminance, and in case of a negative peak of theestimated IF carrier and a high luminance, or the shift (sh) is negativein case of a negative peak of the estimated IF carrier and a lowluminance, and in case of a positive peak of the estimated IF carrierand a high luminance.
 13. Method according to claim 12, wherein thecompressed transport sample (CTS) is in the range [A . . . B] wrappedaround sum of the clipped prediction error ((e_(enc))_(c)) and a mappedvalue m(y) with y being either the prediction {circumflex over(x)}_(enc) or the sum of the prediction {circumflex over (x)}_(enc) andthe shift sh, and that the said mapped value m(y) is in a range [D . . .E], E and D being integers and E−D being equal to or smaller than2^((N−C)) ⁻¹, the mapping being either a uniform or a non-uniformmapping, the non-uniform mapping being such that the quantization isfine for prediction values corresponding to small input amplitudes andrough for prediction values corresponding to big amplitudes of the firstdigital signal.
 14. Transmitting apparatus wherein a digitizedtelevision IF signal is transformed into a transport channel bit-stream(TCBS) for transmission of the said digitized television IF signal froma first location (FL) to one or more second locations (SL), comprising:a splitter (SP) for splitting a sample (S1) of the said digitizedtelevision IF signal into N most significant bits (MSB) and M−N leastsignificant bits (LSB). an encoder DPCM-core (D1) for compression of theN most significant bits (MSC) of a sample (S1) into a N−C bit compressedtransport sample (CTS), generating a clipped prediction error((e_(enc))_(c)). an output for the transport channel bit-stream (TCBS).a first location clipping detector (BSC1 or BSC12) which generates afirst location PCM-bit substitution control signal (SC1) indicating whatis to be transmitted as residual transport sample (RTS), either the M−Nleast significant bits (LSB) of the sample S1 of the first digitizedtelevision IF signal, or a substitution value (CE) being a firstfunction of both the clipping error corresponding to the said sample S1of the first digitized television IF signal, and the M−N ISB's of thesaid sample S1 of the first digitized television IF signal, and thefirst location substitutor (BS1) which substitutes the M−N leastsignificant bits (LSB) by a substitution value (CE), depending on thevalue of the first location PCM-bit substitution control signal (SC1).15. Transmitting apparatus according to claim 14, further comprising: aprediction mapper (PM1) for generating a mapped prediction m({circumflexover (x)}_(enc)) from an encoder prediction ({circumflex over(x)}_(enc)) from the encoder CPCM-core (D1), and an adder (ADD 12) whichadds the mapped prediction (m({circumflex over (x)}_(enc)) and theclipped prediction error ((e_(enc))_(c)) and then wraps around theresult of the addition, thus obtaining a compressed transport sample(CTS), and wherein the transmitting apparatus includes at least one of(A) and (B): (A) the encoder DPCM-core (Dl) comprises means to clip theprediction errors to a range equal to or smaller than 2^((N−C)) ⁻¹, and(B) the prediction mapper (PM1) comprises means for a uniform or anon-uniform mapping.
 16. Transmitting apparatus wherein a digitizedtelevision IF signal is transformed into a transport channel bit-stream(TCBS) for transmission of the said digitized television IF signal froma first location (FL) to one or more second locations (SL), comprising:an encoder DPCM-core (D1) for compression of a sample S1 of the firstdigitized television IF signal, represented by N bits, into a N−C-bitcompressed transport sample (CTS), generating a prediction (ê_(enc)) anda clipped prediction error ((e_(enc))_(c), an output for the transportchannel bit-stream (TCBS), a prediction mapper (PM1) for generating amapped prediction (m({circumflex over (x)}_(enc))) from the prediction({circumflex over (x)}_(enc)) from the encoder DPCM-core (D1), and anadder (ADD12) which adds the mapped prediction (m({circumflex over(x)}_(enc))) and the clipped prediction error ((e_(enc))_(c)), and thenwraps around the result of the addition, thus obtaining a compressedtransport sample (CTS), and wherein the transmitting apparatus includesat least one of (A) and (B): (A) the encoder DPCM-core (D1) comprisesmeans to clip the prediction errors to a range equal to or smaller than2^((N−C)) ⁻¹, and (B) the prediction mapper (PM1) comprises means for auniform or a non-uniform mapping.
 17. Transmitting apparatus accordingto claim 14, further comprising: a phase-locked loop (PLL1) whichestimates the phase (Φ_(enc)) of the IF carrier of the digitizedtelevision IF signal, based on a locally decoded television IF signal({tilde over (x)}_(enc)) from the encoder DPCM-core (D1), a luminanceestimator (LUE1) which estimates the luminance of the video signalcontained in the digitized television IF signal, based on the decodedtelevision IF signal ({tilde over (x)}_(enc)) and on the estimated phase(Φ_(enc)) of the IF carrier, resulting in an estimated luminance(L_(enc)), and a shift estimator (SHE1) which estimates a shift(sh_(enc)), based on the estimated phase (Φ_(enc)) of the IF carrier andon the estimated luminance (L_(enc)), wherein the encoder DPCM-corecomprises means to clip the prediction error (e_(enc)) to a range whichis shifted over a shift (Sh_(enc)).
 18. Transmitting apparatus wherein adigitized television IF signal is transformed into a transport channelbit-stream (TCBS) for transmission of the said digitized television IFsignal from a first location (FL) to one or more second locations (SL),comprising: an encoder DPCM-core (D1) for compression of a sample (S1)of the first digitized television IF signal, represented by N bits, intoa N−C-bit compressed transport sample (CTS), generating a prediction(ê_(enc)) and a clipped prediction error (e_(enc))_(c)) an output forthe transport channel bit-stream (TCBS), a phase-locked loop (PLL1)which estimates the phase (Φ_(enc)) of the IF carrier of the digitizedtelevision IF signal, based on a locally decoded television IF signal({tilde over (x)}_(enc)) from the encoder DPCM-core (D1), a luminanceestimator (LUE1) which estimates the luminance of the video signalcontained in the digitized television IF signal, based on the decodedtelevision IF signal ({tilde over (x)}_(enc)) and on the estimated phase(Φ_(enc)) of the IF carrier, resulting in an estimated luminance(L_(enc)), and a shift estimator (SHE1) which estimates a shift(sh_(enc)), based on the estimated phase (Φ_(enc)) of the IF carrier andon the estimated luminance (L_(enc)), wherein the encoder DPCM-corecomprises means to clip the prediction error (e_(enc)) to a range whichis shifted over a shift (sh_(enc)).
 19. Transmitting apparatus accordingto claim 18, further comprising: a prediction mapper (PM1) forgenerating a mapped value m(y) of either the encoder predictiony={circumflex over (x)}_(enc) from the encoder DPCM-core (D1) or the sumy={circumflex over (x)}_(enc)+sh of the encoder prediction ê_(enc) andthe clip range shift sh, and an adder (ADD 12) which adds the mappedvalue (m(y)) and the clipped prediction error ((e_(enc))_(c)) and thenwraps around the result of the addition, thus obtaining a compressedtransport sample (CTS), and wherein the transmitting apparatus includesat least one of (A) and (B): (A) the encoder DPCM-core (D1) comprisesmeans to clip the prediction errors to a range equal to or smaller than2_((N−C))−1, and (B) the prediction mapper (PM1) comprises means for auniform or non-uniform mapping.
 20. Transmitting apparatus according toclaim 17, further comprising: a prediction mapper (PM1) for generating amapped value m(y) of either the encoder prediction y={circumflex over(x)}_(enc) from the encoder DPCM-core (D1) or the sum y={circumflex over(x)}_(enc)+sh of the encoder prediction {circumflex over (x)}_(enc) andthe clip range shift sh, and an adder (ADD 12) which adds the mappedvalue (m(y)) and the clipped prediction error (e_(enc))_(c), and thenwraps around the result of the addition, thus obtaining a compressedtransport sample (CTS), and the transmitting apparatus includes at leastone of (A) and (B) (A) the encoder DPCM-core (Dl) comprises means toclip the prediction errors to a range equal to or smaller than2_((N−C))−1, and (B) the prediction mapper (PM 1) comprises means for auniform or non-uniform mapping.
 21. Receiving apparatus wherein atransport channel bit-stream (TCBS), which contains a first digitizedtelevision IF signal represented by transport samples composed of atleast a N−C-bits compressed transport sample (CTS) and a (M−N)-bitsresidual transport sample (RTS) and which is either obtained throughcompression or is transmitted, is transformed into a second digitizedtelevision IF signal, comprising: an input for the transport channelbit-stream (TCBS). a decoder DPCM-core (D2) for decompressing the N−Cbits of a compressed transport sample (CTS) to N MSB's of an outputsample (S2), a combiner (CB) for combining the M−N LSB's and N MSB's ofa sample to an output sample (S2). a second location clipping detector9BSC2) which generates a second location PCM-bit substitution controlsignal (SC2), indicating what is to be selected as M−N least significantbits of the output sample (S2), and a sign signal (SGN), being the signbit of the transmitted transformed clipping error, a second locationsubstitutor (BS2) which switches between the received M−N LSB's from theresidual transport sample (RTS) and a replacement according to thesecond location PCM-bit substitution control signal (SC2), and an MSBcorrector (COR) which adds to or subtracts from the result of adderADD21 in the decoder DPCM-core (D2) and the output value of mapping thereceived transport sample (RTS) by mans of a second function, accordingto the second location PCM-bit substitution control signal (SC2) and thesign signal (SGN).
 22. Receiving apparatus according to claim 21 furthercomprising: a prediction mapper (PM2) for generating a mapped prediction(m({circumflex over (x)}_(dec))) from a decoder prediction ({circumflexover (x)}_(dec)) from the decoder DPCN-core (D2), whereby the predictionmapper (PM2) comprises means for one or a uniform mapping and anon-uniform mapping, and a subtractor (SUB21) which subtracts the mappedprediction (m({circumflex over (x)}_(dec))) from the sample of thecompressed transport stream (CTS) and which then does a wrap-around ofthe result of the subtraction, obtaining to a clipped prediction error((e_(dec))_(c)).
 23. Receiving apparatus wherein a transport channelbit-stream (TCBS), containing a first digitized television IF signalrepresented by transport samples composed of at least a (N−C)-bitscompressed transport sample (CTS) and either being compressed ortransmitted, is transformed into a second digitized television IFsignal, comprising. an input for the transport channel bit-stream(TCBS), a decoder CPCM-core (D2) for decompressing the N−C bits of acompressed transport sample (CTS) to N bits of an output sample (S2), aprediction mapper (PM2) for generating a mapped prediction(m({circumflex over (x)}_(dec))) from a decoder prediction (x_(dec))from the decoder DPCM-core (D2), whereby the prediction mapper (PM2)comprises means for one of a uniform mapping and a non-uniform mapping,and a subtractor (SUB21) which subtracts the mapped prediction(m({circumflex over (x)}_(dec))) from the sample of the compressedtransport stream (CTS) and which then does a wrap-around of the resultof the subtraction, obtaining to a clipped prediction error((e_(dec))_(c).
 24. Receiving apparatus according to claim 21 furthercomprising: a phase-locked loop (PLL2) which estimates a phase (φ_(dec))of the IF carrier based on a decoded television IF signal ({tilde over(x)}_(dec)) from the decoder DPCM-core (D2), a luminance estimator(LUE2) which estimates the luminance of the video signal contained inthe television IF signal based on the decoded television IF signal({tilde over (x)}_(dec)) and on the said estimated phase (φ_(dec)) ofthe IF carrier, resulting in an estimated luminance (L_(dec)) and ashift estimator (SHE2) which estimates the amount of shift (sh_(dec))based on the said estimated phase (φdec) of the IF carrier and on thesaid estimated luminance (L_(dec)), and wherein said decoder CPCM-core(D2) comprises means to decode prediction errors which have been clippedin the encoder at the corresponding transmitting apparatus to a cliprange which has been shifted there by an amount indicated by a shiftestimator (SHE1).
 25. Receiving apparatus wherein a transport channelbit-stream (TCBS), containing a first digitized television IP signalrepresented by transport samples composed of at least a (N−C)-bitscompressed transport sample (CTS) and either being compressed ortransmitted, is transformed into a second digitized television IFsignal, comprising, an input for the transport channel bit-stream(TCBS), a decoder DPCM-core (D2) for decompressing the N−C bits of asample of the compressed transport stream (CTS) to N bits of an outputsample (S2), a phase-locked loop (FLL2) which estimates a phase(φ_(dec)) of the IF carrier based on a decoded television IF signal({tilde over (x)}_(dec)) from the decoder DPCM-core (D2), a luminanceestimator (LUE2) which estimates the luminance of the video signalcontained in the television IF signal based on the decoded television IFsignal ({tilde over (x)}_(dec)) and on the said estimated phase(φ_(dec)) of the IP carrier, resulting in an estimated luminance(L_(dec)) and a shift estimator (SHE2) which estimates the amount ofshift (sh_(dec)) based on the said estimated phase (φ_(dec)) of the IFcarrier and on the said estimated luminance (L_(dec)). wherein saiddecoder CPCM-core (D2) comprises means to decode prediction errors whichhave been clipped in the encoder at the corresponding transmittingapparatus to a clip range which has been shifted there by an amountindicated by a shift estimator (SHE1).
 26. Receiving apparatus accordingto claim 25, further comprising: a prediction mapper (PM2) forgenerating a mapped value (m(y)) from the decoder predictiony={circumflex over (x)}_(dec) or the sum y={circumflex over(x)}_(dec)+sh of decoder prediction {circumflex over (x)}_(dec) from thedecoder CPCM-core (D2) and the shift amount sh, whereby the predictionmapper (PM2) comprises means for one of a uniform mapping and anon-uniform mapping, and a subtractor (SUB21) which subtracts the mappedvalue (m(y)) from the sample of the compressed transport stream (CTS)and which then does a wrap-around of the result of the subtraction,obtaining so a clipped prediction error ((e_(dec))_(c).
 27. Receivingapparatus according to claim 24, further comprising: a prediction mapper(PM2) for generating a mapped value (m(y)) from one of the decoderprediction y={circumflex over (x)}_(dec) and the sum y={circumflex over(x)}_(dec)+sh of decoder prediction dec from the decoder CPCM-core (D2)and the shift amount sh, whereby the prediction mapper (PM2) comprisesmeans for one of a uniform mapping and a non-uniform mapping, and asubtractor (SUB21) which subtracts the mapped value (m(y)) from thesample of the compressed transport stream (CTS) and which then does awrap-around of the result of the subtraction, obtaining so a clippedprediction error ((e_(dec))_(c).